System for operating DC motors and power converters

ABSTRACT

A system is disclosed for driving a DC motor ( 15 ) under conditions of a controlled average current. An inductive element may be arranged for connection in series with the DC motor. A switch ( 14 ) is preferably coupled to the inductive element for connecting and disconnecting a terminal of the inductive element from the voltage source. A diode may be arranged for connection in parallel with a combination of the inductive element and the DC motor arranged in series, with the appropriate polarity so that current circulating through the inductive element circulates through the diode when the switch disconnects the terminal from the voltage source. A capacitor is arranged for connection in parallel with the motor, for limiting a resulting voltage over the motor or for storing charge depending on the embodiment of the invention. A device for measuring a current through the motor is provided, and a device ( 13 ) for controlling operation of the switch dependent upon the measured current in the motor is also provided. An airflow apparatus is also disclosed.

FIELD OF THE INVENTION

The present invention relates generally to motors and power suppliesand, in particular, systems for operating a direct-current (DC) motor orpower converters.

BACKGROUND OF THE INVENTION

In the treatment of sleep apnoea and other respiratory disorders, apositive air pressure is used applied to the patient airway. Theequipment used is known as an airflow generator.

A method used to generate air pressure is shown in FIG. 1. A brushlessDC motor (16) is used to drive a turbine or blower (15). The turbine(15) generates the air flow for the patient. The brushless DC motorcontroller (14), in conjunction with the control electronics of the flowgenerator (13), receive power from a power supply (12) that is connectedto the AC main through a filter (11). Sometimes the filter is built intothe power supply itself. Control signals are sent from the controlelectronics (13) to the brushless DC motor controller (14), so the speedof the motor (16) can be controlled.

The pressure and the amount of air delivered depend on the speed of theturbine. In some types of equipment, pressure and flow sensors are usedto monitor these variables and change the speed of the motor to achievethe desired effect. Also, in some cases, the speed of the motor ischanged, alternating between a high and a low value, either in responseto the patient respiration or as part of an automatic cycle. Equipmentperforming in this way is known as bi-level devices.

In FIG. 1, a single power supply (12) provides power to both, the motordriving circuits (14) and the control electronics (13).

A brushless DC motor, or BLDCM, is a DC motor with an electroniccommutator. FIG. 2 shows a block diagram of one type of permanent magnetbrushless DC motor along with its electronic commutator.

The driving electronics consist of a logic circuit (22) that controls aset of electronic switches (21) that switch power to the motor windings(23) much as the brushes do in a conventional DC motor. Current throughthe windings (23) generates forces in the rotating magnets (24), causingthe rotor of the motor to spin. The switches (21) can connect the end ofits corresponding winding to either the positive or the negative side ofthe DC voltage source, and also they can leave the winding unconnected.

The logic circuit (22) of the electronic commutator has as an output twocontrol signals per switch, shown in the figure as signal groups SWC1 toSWC3, of two lines each. The motor has hall-effect sensors (25, 26 and27) that are used by the logic circuit (22) to detect the position ofthe rotor and to switch the right waveforms to the windings (23).Typically, the industry uses a three phase motor (three windings) thatis depicted in Y-configuration, for example, but may also be in atriangle configuration.

As the axis of the motor rotates, the motor windings are driven withthree trapezoidal 6-step waveforms. During each step, voltage is appliedto two windings only.

There is also a sensor-less mode of operation, in which a specialcontroller monitors the voltage in the winding that has been leftopen-circuit to read the back-emf generated in the winding as the motoraxis rotates.

In a CPAP application, like the one shown in FIG. 1, the BLDCM (16)takes considerable power especially during its acceleration periods. Ina typical CPAP application, a motor can take around two amps at 24 volts(or more if a 12 volt motor is used), depending on the pressure and flowgenerated, and the particular motor chosen. The electronics necessary toperform the control, however, can be designed so the electronics takeunder 0.1 amps of current at a relatively low voltage. Most of theelectronics can work with 5 volts, while only the pressure and flowsensors may need more, depending on the implementation.

The power supply for the motor (16 in FIG. 1) and the switching sectionof the electronic commutator (21 in FIG. 2) require more relaxedspecifications than the power supply for the control electronics (13 inFIG. 1) or the electronic commutator logic (22 in FIG. 2). A motor is aforgiving load for a power supply. As the motor's mechanicalcharacteristics work as a low pass filter, the motor can tolerate arelatively large ripple voltage. In fact, the ripple can be up to 100%without affecting operation. Furthermore, some applications of motors(e.g., driving a fan or turbine) can tolerate the discontinuous torquethat comes with a discontinuous supply of current.

A brushless-DC-motor-driven ventilation fan shares most of the buildingblocks of an airflow generator for medical applications. The maindifferences are:

-   -   The mechanical design of the turbine or fan itself, since a flow        generator needs to produce more pressure.    -   The ventilation fan, normally, does not interface with flow and        pressure sensors. Thus, the control electronics of a ventilation        fan are simpler and should draw less current.

Regulations like the European Standard EN 60555 and the InternationalStandard IEC 555-2 limit the current harmonic content of mains suppliedequipment. This requirement applies to both the medical application ofthe DC motors and the ventilation fans. Power factor correction must betaken into account for all new designs. Power factor correction can add20 to 30% to the cost of the power supply of equipment (see Ref. 10 inAppendix C). Hence, there can be a relatively substantial saving in thecost of the equipment if the function of power factor correction isintegrated with the DC motor driver for equipment working from the ACmains.

SUMMARY OF THE INVENTION

In accordance with an aspect of the invention, there is provided asystem for driving a direct-current (DC) motor under conditions of acontrolled average current, the system comprising: an inductive elementfor connection in series with the DC motor; a first switch coupled tothe inductive element for connecting and disconnecting a terminal of theinductive element remote from the DC motor to a voltage source; a secondswitch connected in parallel with a combination of the inductive elementand the DC motor arranged in series, controlled so that a currentcirculating through the inductive element circulates through the secondswitch if the first switch disconnects the terminal of the inductiveelement from the voltage source; a capacitor arranged for connection inparallel with the DC motor to limit a resulting voltage over the DCmotor; means for measuring a current through the DC motor; and means forcontrolling operation of the first and second switches dependent uponthe measured current in the DC motor.

In accordance with another aspect of the invention, there is provided asystem for driving a direct-current (DC) motor under conditions ofcontrolled average current, the system comprising: a capacitor arrangedfor connection in parallel with the motor to limit a resulting voltageover the motor, the other terminal of the parallel combination of thecapacitor and the motor being connected to a common terminal; aninductive element connected to the common terminal; a first switchcoupled to the inductive element for connecting and disconnecting aterminal of the inductive element to a voltage source; a second switchconnected in series with the parallel combination of the motor and thecapacitor, and connected to the common node between the first switch andthe inductive element, controlled so that the current circulatingthrough the inductive element circulates through the second switch ifthe first switch disconnects the terminal from the voltage source; meansfor measuring a current through the motor; and means for controllingoperation of the first and second switches dependent upon the measuredcurrent in the motor.

In accordance with yet another aspect of the invention, there isprovided a system for driving a direct-current (DC) motor underconditions of a controlled average current, the system comprising: acapacitor arranged for connection in parallel with the motor to limit aresulting voltage over the motor, a terminal of the capacitor and themotor being connected to a DC voltage source; an inductive elementconnected to a common node of the DC voltage source, the capacitor andthe motor; a first switch coupled to the inductive element forconnecting and disconnecting a terminal of the inductive element to theterminal of the voltage source not connected to the parallel combinationof the capacitor and the motor; a second switch connected in series withthe parallel combination of the motor and the capacitor and connected tothe common node between the first switch and the inductive element,controlled so that the current circulating through the inductive elementcirculates through the second switch if the first switch disconnects theterminal from the voltage source; means for measuring a current throughthe motor; and means for controlling operation of the first and secondswitches dependent upon the measured current in the motor.

In accordance with a further aspect of the invention, there is provideda system for driving a direct-current (DC) motor under conditions of acontrolled average current, a voltage of a DC power supply having alarger or smaller value than a motor nominal voltage, the systemcomprising: an inductive element for connection in series with the DCmotor; an arrangement including a plurality of switches, diodes and amagnetic system, the arrangement being coupled to the inductive elementfor connecting and disconnecting a terminal of the inductive elementremote from the motor to a voltage source, the arrangement beingconfigured as circuit selected from the group consisting of a forwardDC-DC converter, a push-pull DC-DC converter, a half-bridge DC-DCconverter, a diagonal-half bridge DC-DC converter, and a full bridgeDC-DC converter; a capacitor arranged for connection in parallel withthe motor to limit a resulting voltage over the motor; means formeasuring a current through the motor; and means for controllingoperation of the arrangement dependent upon the measured current in themotor.

In accordance with still another aspect of the invention, there isprovided a system for driving a direct-current (DC) motor underconditions of a controlled average current, a voltage of a DC powersupply having a larger or smaller value than a motor nominal voltage,the system comprising: a diode; a magnetic transformer connected inseries with the diode in a circuit arrangement selected from the groupconsisting of a flyback DC-DC converter and a ringing choke DC-DCconverter, the transformer and the diode for connection in series withthe DC motor; a switch coupled to the magnetic transformer and the diodefor connecting and disconnecting a terminal of the magnetic transformerand the diode remote from the motor to a voltage source; a capacitorarranged for connection in parallel with the motor to limit a resultingvoltage over the motor; means for measuring a current through the motor;and means for controlling operation of the switch dependent upon themeasured current in the motor.

In accordance with another aspect of the invention, there is provided asystem for driving a direct-current (DC) motor under conditions of acontrolled average current, a voltage of a DC power supply having alarger or smaller value than a motor nominal voltage, the systemcomprising: an electronic synchronous rectification switch; a magnetictransformer connected in series with the synchronous rectificationswitch in a circuit arrangement selected from the group consisting of aflyback DC-DC converter and a ringing choke DC-DC converter, thetransformer and the synchronous rectification switch being forconnection in series with the DC motor; a switch coupled to the magnetictransformer and the synchronous rectification switch for connecting anddisconnecting a terminal of the magnetic transformer and the synchronousrectification switch remote from the motor to a voltage source; acapacitor arranged for connection in parallel with the motor to limit aresulting voltage over the motor; means for measuring a current throughthe motor; and means for controlling operation of the switch dependentupon the measured current in the motor.

In accordance with yet another aspect of the invention, there isprovided an airflow apparatus, comprising: a brushless DC motor; anelectronic circuit for controlling operation of the brushless DC motor;a power supply for the electronic circuit separate from a power supplyfor the brushless DC motor, the power supply for the electronic circuitbeing adapted to use a voltage resulting from the brushless DC motor inoperation once the resulting voltage reaches a suitable value; and meansfor reducing power to the electronic circuit from the power supply oncethe resulting voltage reaches the suitable value.

In accordance with a further aspect of the invention, there is provideda system for powering a microprocessor based system from a DC voltagehigher than the voltage required by the system to operate, comprising: acapacitor; means to charge the capacitor from the DC voltage with acurrent substantially smaller than the current the microprocessor basedsystem needs to operate; a switch coupled to the capacitor so that theswitch can connect power to the microprocessor based system from thecharge accumulated in the capacitor; means for sensing the voltage inthe capacitor and causing the switch to close once the voltage in thecapacitor reaches a desired value; and means for keeping the switchclosed while the voltage in the capacitor is over a desired value, butless than the value that caused the sensing means to close the switch.

In accordance with another aspect of the invention, there is provided aswitching based alternating current (AC) to direct-current (DC)converter, comprising: a rectifier adapted to be connected to analternating current (AC) mains line; a first capacitor for noisereduction connected in parallel with an output of the rectifier; aninductive element connected to a terminal of the rectifier and the firstcapacitor; a first switch coupled to the inductive element forconnecting and disconnecting a terminal of the inductive element remotefrom the parallel combination of the rectifier output and the firstcapacitor; a second switch connected to the connection node between theinductive element and the first switch, controlled so that the currentcirculating through the inductive element circulates through the secondswitch when the first switch disconnects the inductive element from theparallel combination of the rectifier output and the first capacitor; asecond capacitor for energy storage connected to the terminal of thesecond switch remote from the inductive element, the second capacitorbeing connected in parallel with the serial combination of the secondswitch and the inductive element, the direct-current (DC) output of thealternating current (AC) to direct-current (DC) converter being takenfrom the terminals of the second capacitor; means for sensing a currentthrough the inductive element; means for sensing the voltage across thefirst capacitor; means for sensing the voltage across the secondcapacitor; and a control circuit connected to the first switch tomaintain the voltage across the second capacitor between defined limitsby operating the first switch in a way that a current through theinductive element tracks the waveform of the alternating current linevoltage to cause the AC-to-DC converter to exhibit unity power factor tothe alternating current line.

BRIEF DESCRIPTION OF THE DRAWINGS

A small number of embodiments are described with reference to thedrawings, in which:

FIG. 1 is a block diagram of an airflow generator with a common powersupply for the control electronics and the brushless DC motor driver;

FIG. 2 is a block diagram of a brushless DC motor included for the sakeof clarity;

FIG. 3 is a block diagram of an airflow generator with split powersupply in accordance with an embodiment of the invention;

FIG. 4 is a block diagram of an airflow generator with split powersupply and the power supply for the control electronics circuit, adaptedto use a voltage resulting from the brushless DC motor in operation oncethe resulting voltage reaches a suitable value;

FIG. 5 is a block diagram of a known DC motor driven with controlledconstant average current;

FIG. 6 is a block diagram of a system for driving a direct-current (DC)motor under conditions of controlled average current, from a voltagesource larger than a voltage that the DC motor has under operatingconditions;

FIG. 6A is a set of timing charts of the waveforms present in FIG. 6;

FIG. 6B is a more detailed block diagram of the system disclosed in FIG.6, including timing of the waveforms of one embodiment of the invention;

FIGS. 6C-1 and 6C-2 are lists of the equations that describe thebehavior of the system shown in FIG. 6;

FIGS. 7A to 7L are block diagrams of further embodiments of theinvention relating to the system depicted in FIG. 6;

FIGS. 8A to 8C are block diagrams contrasting embodiments of theinvention and the known system using switched mode power supplies;

FIGS. 9A to 9L are circuit diagrams for detailed implementations of flowgenerators in accordance with an embodiment of the invention;

FIG. 10A is a more detailed conceptual block diagram of an airflowgenerator with an auxiliary power supply and an electronic switch or acurrent summing point to use a voltage resulting from the voltage on thebrushless DC motor while in operation, once the resulting voltagereaches a suitable value;

FIG. 10B is a block diagram of a system with an auxiliary power supplyand the electronic switch;

FIG. 10C is a circuit diagram of a DC-DC converter system to adapt thevoltage developed over the motor to power the controlling electronics inaccordance with FIG. 10A;

FIG. 10D is a block diagram of a system that allows the use of a simpleauxiliary power supply without the need for low power electronics;

FIG. 10E is a circuit diagram of an electronic switch used in FIG. 10D;

FIGS. 11A and 11B are diagrams of an airflow generator for CPAPtreatment implementing an embodiment of the invention;

FIG. 12 is the block diagram of a system that shows an embodiment of theinvention, in the form of a multiple cooling fan controller;

FIG. 13 is a block diagram of a system that shows another embodiment ofthe invention, in the form of a single cooling fan controller;

FIG. 14 is a block diagram of another embodiment of the invention;

FIG. 15 is a block diagram of the embodiment of the invention shown inFIG. 7J with accompanying analysis;

FIG. 16 is a block diagram of a topology for a power factor correctedswitch-mode AC-to-DC converter; and

FIGS. 17A to 17F are diagrams of embodiments of the invention shown inFIG. 16.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Systems for driving a direct-current (DC) motor under conditions of acontrolled average current, airflow apparatuses, and systems forpowering a microprocessor based system from a DC voltage higher than thevoltage required by the system to operate are described. Also describedis a typology for power factor correction.

Embodiments of the invention provide a simple, less costly system thatis obtained if the motor is made to work as directly as possible fromthe AC main while powering the control electronics from a smaller,simpler well-regulated power supply. However, the BLDCMs available tothe manufacturers of CPAP flow generators do not operate from thevoltage available after the rectification of the AC main. Furthersimplification, integration and versatility can be obtained if the samecontrol electronics can look after the control and mode of operation ofthe power supply itself, and the implementation of the functionality ofthe CPAP flow generator.

In other embodiments of the invention, a microprocessor can provide theintelligence necessary for controlling the functionality of the CPAPflow generator, to also control the supply of power to the BLDCMcontroller, while still providing extra functionality like power factorcorrection, and modulation of the switching frequencies used in thepower supply in order to facilitate meeting the EMI directives.

The advantage of powering the motor as directly as possible from the ACmain brings one design for a portable airflow generator for CPAPtreatment that in terms of simplicity uses a BLDCM that is connecteddirectly to a pulsating full wave rectified AC main. The concept isshown in FIG. 3. The control electronics (33) in FIG. 3 are powered fromthe power supply (32) that takes the power from the AC mains through afilter (31). The brushless DC motor, BLDCM (35) that drives the turbine(36) is controlled or driven from the control electronics (34) thattakes power from the AC mains directly through the filter (31). Theelectronic commutator of the BLDCM is included in (34), while the motorsensors that should connect to (34) are not shown for sake ofsimplicity.

As all the BLDCMs (35) suitable to build a portable airflow generatorfor CPAP in the market would not stand the main voltage directly appliedto its windings, alternative methods for driving the BLDCM must be used.

The unavailability of brushless DC motors suitable for use in CPAP flowgenerators that can work at rectified AC main levels has motivated theembodiments of the invention, which can be seen conceptually in FIGS. 4,6 and 7.

While the embodiments of the invention are directed to motors in flowgenerators, it will be appreciated by those skilled in the art that theembodiments of the invention are not limited to such applications.Another application of the invention is BLDCM driven cooling fans, forexample.

The power supply that the motor and the electronic commutator requirehas far more relaxed specifications than the power supply for thecontrol electronics. A motor is a forgiving load for a power supply. Themechanical characteristics of the motor work as a low pass filter, sothe motor can tolerate a relatively large ripple voltage. In fact, theripple can be up to 100% without affecting operation. If the controlledvariable is the motor current, and the electronic commutator is notpulse width modulated, the voltage over the motor's electroniccommutator is the result of the back-emf (electromotive force) generatedin the windings themselves as a result of the motion. This voltage isapproximately a linear function of the motor speed and can be used toestimate the speed of the motor.

There are several advantages in controlling the current through a DCmotor rather than the voltage over that motor. For instance, it is wellknown that by controlling the current driving a DC motor, rather thanthe voltage over the windings, the electrical time constant formed bythe resistance and the inductance of the windings is not part of thecontrol loop dynamics, making the system more responsive. Also, the samecircuitry that is used to control the current can be used simultaneouslyto detect malfunctions due to excess current, yielding a simpler,cheaper circuit. Finally, if the current through the DC motor iscontrolled, conditions of blocked rotor are easier to handle, as thereis no excess current taken from the power source.

FIG. 5 shows a circuit for driving a DC motor (including a BLDCM withits electronic commutator) under conditions of a controlled averagecurrent.

The switch (52) closes and opens as commanded by the pulse widthmodulator (56). When the switch (52) is closed, DC voltage from thesource (51) is applied to the motor (54), and a current flows throughthe inductance (the motor windings) embedded in the motor (L_(M)). Whenthe switch opens, the current flows through the diode (53). The diode(53) is usually called a “freewheeling diode” or a “catch diode”. Acurrent sensor (55), typically a resistor of small value, is used tomeasure the current. Block (57) represents either an error amplifier, ora comparator, depending on the type of control system used. Block (57)compares the measured current with the desired or set current passingthe information, as control signal, to a control system (58), which inturn controls the switch (52).

This type of known system and its variations are not only applied to DCmotors but to other types of electromechanical devices with an embeddedinductance. Examples of these devices are other types of motors, likestepper motors and even solenoids.

The necessary semiconductor circuitry to perform the functions in FIG. 5can be found in integrated circuits well known to those skilled in theart (e.g., see Refs. 5 and 19).

The details of the implementation of FIG. 5 can be found in:

Reference 1, chapter 7, pages 155-181.

Reference 3, chapter 11, pages 286-308.

Reference 4, FIGS. 7 and 8 (technique used with other inductivedevices).

References 5, 6 and 13-19.

Reference 24, pages 3-51 to 3-70 and 5-6 to 5-9.

A limitation of the current state of the art as shown in FIG. 5 is thatwhen the switch (52) closes, the full value of the DC voltage of thesource (51) is applied to the motor (54). Hence, the motor (54) must bespecified accordingly. A motor designed to be driven from a nominalvoltage of 24 volts DC is not, generally, expected to operate from the337 volts DC that results from rectifying a 240 VAC main.

A method for driving a motor with controlled constant average currentthrough the motor windings, while taking the power from a voltage sourcelarger than the maximum voltage source possible for the motor, isdisclosed in FIG. 6.

In the circuit disclosed in FIG. 6, a switch (62) connects the DCvoltage from the power source (61) not to the motor (66) directly, butto an inductive element (64) not present in the circuit shown in FIG. 5.A capacitor (65) is added in parallel with the motor (66).

Referring to FIG. 6, the electronics controlling the operation of thecircuit (68) command a pulse width modulator (69). The object of thecontrol system (68) is to keep the current though the motor (66) asclose as possible to the desired motor current command received fromoutside the system. The control system (68) receives the informationfrom the current sensor (67) and the desired motor current as commandedfrom outside the system, and applies control algorithms or signalprocessing to obtain an output that commands the pulse width modulator(69). The operation of the control system (68) can also be described interms of a comparison of the current through the sensor (67) with thedesired motor current, as commanded from outside the system, and thegeneration of control signals for the pulse width modulator (69) inaccordance with the results of the comparison in order to achieve theobject described above.

The pulse width modulator (69) is coupled to the switch (62) and canmake the switch open and close. The switch (62) opens and closesrepeatedly, as part of a pulse-width modulation scheme generated in thepulse width modulator (69) to obtain the desired current in the motor(66). Every time the switch (62) is closed, the current in the inductiveelement (64) grows in proportion to the voltage difference appliedacross the inductive element (64) and the amount of time the switch isclosed.

When the switch is open, the inductive element (64) keeps the currentflowing through the motor (66) and the capacitor (65), as the circuitcloses through the diode (63). The diode (63) is performing a similarfunction to what is normally referred to as a catch diode or afreewheeling diode. Current sensor (67) senses a current similar to thecurrent in the inductive element (64).

Generally, the capacitor (65) limits the resulting voltage over themotor, if the resulting voltage exceeds specifications for the motor dueto a relationship between the impedance of the inductive element and theimpedance of the motor. However, even if the voltage across the motorresulting from the inductive divider formed by the inductance of themotor and the inductance (64) is within the specification of the motor,the capacitor (65) is needed to take the current that the inductance(64) is forcing into the motor, during the relatively small period oftime in which the electronic commutator of the motor in a brushless DCmotor (or, sometimes, the brushes in a conventional DC motor) may beopen.

FIG. 6B shows a detailed block diagram of one possible implementation ofthe control system (68) and the pulse width modulator (69). The controlsystem and the pulse width modulator shown in FIG. 6B are a popularchoice with integrated circuits. The motor (66) of FIG. 6 has beenreplaced graphically by its electrical model, comprising the inductanceof the windings (L_(M)), the resistance of the commutator and thewinding R_(M), and the counter electromotive force Vemf, represented asa voltage source in series with L_(M). All the elements of FIG. 6B, withnumbers similar to the numbers used in FIG. 6, have similarfunctionality.

The capacitor (65) is shown as an ideal capacitor (65B) and itseffective series resistance (ESR), shown as a resistor labelled R_(ESR),in series with (65B).

A comparator (6B1) compares the current information in the form of avoltage from the sensing resistor (67), that is, the voltage V_(RS),with the set point, or desired motor current information, the voltageV_(DMC).

The cycle of the pulse width modulator starts periodically, when a pulsefrom the oscillator (6B3) sets the set-reset latch (6B2), causing theoutput “Q” to go high. The output “Q” of the set-reset latch (6B2) isconnected to the switch (62). The output “Q” is labelled SWC, fromswitch control. A high in SWC causes the switch (62) to close. Currentflows through the inductance (64), the capacitor (65), the motor (66)and the sensing resistor (67). When the current sensed in (67) is largerthan the set point value (V_(DMC)), the output of the comparator (6B1)is high, and the set-reset latch (6B2) is reset.

With the reset input active, the output “Q” (also labelled S_(WC))changes into low, and the switch (62) opens. The higher the value of thesignal V_(DMC), the higher the current is allowed to grow in theinductance (64), and the higher the average current through the motoris. Also, the time the switch is on (T_(ON)) grows.

The five-chart timing diagram at the bottom of FIG. 6B shows the timingof some waveforms in the circuit:

OSCOUT: Output of the oscillator (6B3). T is the period of the PWMsignal. SWC: Control input to the switch (62), output “Q” of the S-Rlatch (6B2). V_(RS): Voltage as a result of the current flow in thesensing resistor (67). I_(L): Current through the inductance (64).RESET: The output of comparator (6B1), and the reset input of the S-Rlatch (6B2).

The type of control system and pulse width modulator shown in FIG. 6B isonly one example. The control schemes used with the circuit shown inFIG. 5 can be used in the control system (68) and the pulse widthmodulator (69) of FIG. 6. In fact, the circuit disclosed in FIG. 6 canbe implemented with some of the integrated circuits that are currentlyused to implement the circuit disclosed in FIG. 5. An example of this isin the Reference 13.

Reference 13 uses an integrated circuit (UC3842) belonging to the samefamily (similar type of control system and pulse width modulator) as theintegrated circuit (UC2845) used in one of the embodiments of theinvention depicted in FIG. 9H. Reference 23 is the data sheet for theUC3842.

This means that the pulse-width modulation scheme used can be any of thetypes currently used to control the current through relay coils of motorwindings, when the available voltage is compatible with the maximumoperating voltage through the windings. It must be noted, however, thatwith the circuit shown in FIG. 5, there is no need for a capacitor inparallel with the motor and there is also no need for any inductanceexternally connected in series with the motor winding.

The chosen value for capacitor (65) is not critical but cannot be toosmall. If at the frequency of operation of the pulse width modulator(69) the impedance of the capacitor (65) is negligible in comparisonwith the impedance of the motor (66), the value of the current throughthe inductive element does not depend on the characteristics of themotor (66).

Capacitor (65) may be chosen large enough to render the system formed bythe capacitor (65) coupled with the impedance of the motor and theinductance (64) at least slightly over-damped. However, the resultingcapacitor can be big. It is not necessary, if the rate of change of thecontrolled average current is limited.

Because of the current feedback system used in FIG. 6B to keep thecurrent through the motor (66) constant, independently of the motorvoltage, the inductive element (64) does not resonate with capacitor(65). Consequently, any ringing in the voltage between the motor (66)and the common terminal has a frequency (f_(Mring)) given by:f _(Mring)=1/(2π√(L _(M) *C))where L_(M) is the inductance of the windings of the motor (see FIG.6B).

Also, if R_(ESR) is assumed to be negligible in comparison with R_(M),the condition to choose the value of capacitor (65), C, for the outputcircuit in FIG. 6B to be over-damped is:C>L _(M)*4/R _(M) ².

Capacitor (65) has to have a low ESR and good pulse characteristics. Thetype of capacitors used as filters in switched mode power supplies is asuitable choice.

In some of the embodiments of the invention, the capacitor in parallelwith the motor is also used to accumulate charge, making it possible tohave different current through the inductive element and the motor.

Sometimes, with the known systems (FIG. 5), a small value capacitor,typically a ceramic high frequency capacitor, is used to filtercommutation noise in DC motors to improve EMI compliance. This capacitorcannot be confused with capacitor (65).

If the noise filtering capacitor of the current state of the art isremoved, the motor continues to operate. However, this only happens withthe embodiment of the invention, if the inductance of the motor has avalue such that the inductive divider formed with the inductive element(64) yields a resulting voltage compatible with the safe operation ofthe motor. With the capacitor (65) in the circuit, the impedance of themotor conducts only DC current in steady state, and hence is not part ofthe dynamic of the system controlling the current. With some systems, itwould be an advantage not to have current at the frequency of operationof the PWM (69) travelling through the wires to connect to the motor(66), especially if the frequency is high, to have a physically smallinductive element (64).

Some of the pulse-width modulation schemes known that can be used forthe circuit disclosed in FIG. 6 are listed below:

1. Hysteresis PWM

-   -   Cycle by cycle current control. A hysteresis band is set around        the desired current level. The top of the hysteresis band        determines when the switch is turned on and conversely the        bottom of the band determines when to turn the switch off.        Ripple current is determined by the difference between the        levels and the speed of reaction of the circuitry. Must be able        to sense the current during the whole cycle. Frequency is        determined by the value of the currents, the value of the DC        voltage source and the value of the energy (current) storing        inductance.        2. Clocked turn-ON, constant frequency PWM    -   Cycle by cycle current control. A periodic signal (clock) starts        the cycle by turning the switch on. At a desired current level        the switch is turned off. The frequency is determined by the        frequency of the periodic signal. It is necessary to measure the        current while the switch is on, and not during the whole cycle.        It is the most commonly used PWM in systems with current mode        control. It is the system shown in FIG. 6B, used in the IC        chosen for the detailed implementation shown in FIG. 9H (UC2845)        and implemented with small scale integrated circuits in FIG.        11B.        3. Clocked turn-OFF, constant frequency PWM    -   Cycle by cycle current control. Like the type 2 above, but the        periodic signal turns the switch off, to be turned on when the        current has dropped under a set value. It is not necessary to        measure the current during the whole cycle. The frequency is        determined by the frequency of the periodic signal.        4. Two current levels PWM (also called triangle PWM)    -   Cycle by cycle current control. One current level to turn the        switch on, another to turn the switch off. Ripple current        determined by the difference between the levels and the speed of        reaction of the circuitry. Must be able to sense the current        during the whole cycle. Frequency is determined by the value of        the currents, the DC voltage source and the value of the energy        (current) storing inductance.        5. Constant OFF time PWM    -   Cycle by cycle or average current control. The cycle starts by        turning the switch on. When the current reaches a pre-set level,        the switch is turned off and stays off for a fixed amount of        time, to complete the cycle. At the end of the fixed period of        time, the switch is turned on again, starting a new cycle. It is        necessary to measure the current while the switch is on, and not        during the whole cycle. Frequency is determined by the value of        the current, the value of the DC voltage source, the value of        the energy (current) storing inductance and the fixed time off        of the switch. If the average current over a number of cycles is        controlled, the on time is made to vary as a result of the        difference between the desired current and the actual current.        6. Constant ON time, average current PWM    -   The switch is closed for a fixed period of time. At the end of        the fixed period, there is a variable delay, and the cycle        starts again. The frequency is variable. The average current        over a number of cycles is controlled, rather than during each        cycle. The off time is made to vary as a result of the        difference between the desired current and the actual current.        7. Constant frequency, variable duty cycle PWM    -   The duty cycle of a square wave is made to vary as a result of        the difference between the desired current and the actual        current. The average current over a number of cycles is        controlled, rather than during each cycle. It is the most widely        used choice in voltage control mode switch-mode power supply        integrated circuits. It is also available as a peripheral in        microcontrollers.        8. All variable (On time, Off time and the Frequency)    -   This method is described in Reference 7 at page 64.

A more detailed description of the operation of the pulse-widthmodulation schemes above can be found in:

Reference 1, pages 174-179.

Reference 3, pages 64, 93, 104-145, 249 and 297.

Reference 12, pages 70-75.

FIG. 6A shows the idealized waveforms in the system depicted in FIG. 6.The names in the charts are referencing points in the embodiment of FIG.6B. However, as only two cycles are shown, any of the above-listedpulse-width modulation schemes give similar waveforms. Obviously, if alarge enough number of cycles would be considered, different pulse widthmodulators show different timing, since for instance some change thefrequency of operation. However, there always is a “T_(ON)” and a“T_(OFF)”, the analysis can be made in a similar way, the results willbe similar, and with all of them a similar value of average current canbe obtained.

In FIG. 6A, the waveform in the charts, are:

V_(A): Voltage between point A in FIG. 6B and ground. V_(FD): Forwardvoltage diode (63). V_(S): Voltage across the switch (62). V_(DSW):Voltage-drop across the switch (62) when the current I_(L) flows. V_(L):Voltage across the inductive element (64). V_(LP): Peak value of V_(L).V_(LM): Minimum value of V_(L). I_(L): Instantaneous current across theinductive element (64). Δ_(IL): Difference between the maximum and theminimum values of I_(L). I_(LPK): Peak value of I_(L). I_(AVERAGE):Average value of I_(L). Is similar to the current through the motorI_(M). T: Period of the pulse width modulator (It can be variable).I_(M): Motor current, the variable that is controlled, by changing thePWM. I_(C): Current through the capacitor (65). T_(ON): Period of timethe switch (62) is closed, SWC must be high. T_(OFF): Period of time theswitch (62) is open, SWC must be low. V_(C): Voltage over (C65B) thecapacitive part of the model of the real capacitor (65). V_(ESR):Voltage over R_(ESR) the resistive part of the model of real capacitor(65). V_(M): Voltage as indicated in FIG. (6B). SWC: Control input tothe switch (62). A high value causes the (62) to close. I_(S): Currentthrough the switch (62). I_(D): Current though the diode (63).

For all the charts, the independent variable is the time.

Some of the charts in FIG. 6A have the dependent variable not in scale.This was thought to be a better choice than letting the small value ofsome variables like V_(C), V_(DSW) and V_(FD) hide their existence inthe charts. If the power source (61) in FIG. 6 is rectified AC mainvoltage, the value of V_(LP), in the chart labelled V_(L), has to bedrawn out of scale, as shown in the chart.

FIGS. 6C-1 and 6C-2 have the equations describing the operation of thecircuit disclosed in FIG. 6, using the embodiment shown in FIG. 6B. Theequations hold true even if a different pulse-width modulation scheme isused. With some types of pulse width modulator, the frequency is notconstant. The equations, however, are correct for each value of theparameters, once steady state is reached. While most of the variableshave already been listed above, the exceptions are:

V_(DC): Value of the voltage of the DC power source (61). R_(M): Theresistance of the windings and commutator of the motor (66). Vemf:Counter electromotive force of motor (66). It is the voltage induced inthe windings as the motor axis rotates. Kv: DC motor constant. Thevoltage of Vemf induced for unit of angular speed (Volt * sec). ω:Angular speed of motor axis (1/sec). V_(RS): Voltage that is the resultof the current been measure using the sensing resistor (67). R_(S):Value of the sensing resistor used to measure the current. The resistorvalue has to be negligible in comparison with the resistance of themotor R_(M). R_(ESR): Effective serics resistance of the real capacitor(65). I_(LMIN): Minimum value of the current through the inductance(64). F: Frequency of the pulse width modulator F = 1/T.

The analysis used the fact that as the impedance of the capacitor (65)at the frequency of operation F is negligible in comparison with theimpedance of the motor (66), all the AC components (the terms containinga multiple of the angular frequency, in the Fourier series expansion) ofthe pulse width modulated waveform can be considered to circulatethrough the capacitor. That sum of terms is what makes the currentI_(C). The rest of the current through the inductive element (64)circulates through the motor (66). The current through the realcapacitor (65) causes a voltage drop, due to the impedance of thecapacitive (C) component of the capacitor (65) and also its effectiveseries resistance R_(ESR). For the analysis to be accurate, this voltagedrop has to be negligible in comparison with the motor voltage V_(M).

Equation (3) in FIG. 6C-1 shows this assumption, as it states that themotor voltage is approximately equal to I_(M)*R_(M)+Vemf.

The equations in FIGS. 6C-1 result from applying Kirchhoff's laws toFIG. 6B.

It is also assumed that the inductive element (64) has no seriesresistor associated with the inductive element. Hence the equation (15)is true and results in equations (16), (17) and (18), of FIGS. 6C-1,when the switch (62) is closed, and in equation (19), of FIGS. 6C-2,when the switch is open and the current is circulating through the diode(63).

Equation (20) shows, as it can be seen in the chart I_(L) of FIG. 6A,that in steady state, the net change of current through the inductiveelement per cycle is zero. That is, the current grows when the switch isclosed and is equal to the current drop when the switch is open.

Equation (21) eliminates the voltage drop over the switch (62) (V_(S))and the forward voltage drop over the diode (63) (V_(FD)), as they arenegligible in comparison with the terms they are subtracted from oradded to.

Equation (22) follows from equation (21) and gives the relationship thatthe duty cycle (T_(ON)/T) of the pulse width modulator has with themotor voltage, as a consequence of the motor current.

This system differs from a current driven buck converter in that thevoltage over the load is not a regulated variable, but it is the resultof the effect of the current itself in the load. In a DC motor, thevoltage in the load is related to the back electromotive force (Vemf)induced in the motor windings as a result of the motion of the windingsin the magnetic fields inside the motor. In fact, given the typicalvalues of resistor RM, and the voltage drop caused on it by the motorcurrent, VM can be used as an estimate of the speed of the motor.

The differences with the current fed buck converter and the well-knowndown converter switched mode power supply are:

DF_1. Under normal operation, the voltage in the load is not taken intoaccount by the control loop, while in a switched mode power supply, isthe controlled variable. With a DC motor, its voltage depends on theback emf. Hence, the voltage depends on the speed of rotation of themotor axis. DF_2. In a switched mode power supply, the capacitor inparallel with the load filters the ripple voltage and may provide thecurrent to the load when the energy accumulated in the inductance dropsto zero before the end of the period of the pulse width modulator (whatis normally called discontinuous operation). In the embodiment of theinvention however, if the current through the inductance drops to zero,the inertia of the motor keeps the motor rotating, and the voltage overthe motor is kept by the electromotive force induced in the winding asthe motor axis rotates.

The analysis of the system of FIG. 6B continues in FIGS. 6C-2 for the ACcomponent of the voltage in V_(M) due to the voltage drop over thecapacitor (65) while in steady state.

The graph in FIGS. 6C-2 is a detailed enlargement of the chart labelledI_(C) of FIG. 6A. It has to be examined keeping in mind that the currentthrough capacitor (65) is similar to the current through the inductiveelement (64), where the average current I_(AVERAGE) has been subtracted.The area shaded in the drawing, labelled ΔQ, is the total amount ofcharge that is transported during one-half of the cycle of the pulsewidth modulator, in or out of the capacitor (65), from the inductiveelement (64).

Equation (23) in FIGS. 6C-2 is the area of the triangle that has beenshaded in the drawing (half the base multiplied by the height). Equation24 in FIGS. 6C-2 shows the relationship between the current variationsthrough the inductive element (64) and the peak voltage in thecapacitive component (C) of the model of capacitor (65).

To get the voltage over capacitor (65), the voltage drop over theresistive component R_(ESR) of the model of capacitor (65) must beadded, as is shown in equation (25) of FIGS. 6C-2.

If the typical figures for the variables in equation (25) are replaced,the result of equation (26) of FIGS. 6C-2 is reached, validating theoriginal assumption regarding V_(M) in equation (3) of FIGS. 6C-1.

Numerous variations in the implementation of the circuit disclosed canbe made.

The inductance, for instance, can be the inductance of the winding of atransformer. In this case, other windings could produce uncontrolledvoltages that can be used to power electronic circuits. This scheme hasbeen widely used in switched mode power supplies.

Another modification of the system disclosed above is that the sense ofthe current can be changed by changing the polarity of the voltagesource. This can be easily done by replacing the switch in FIG. 6 with abridge circuit as is commonly done in four-quadrant control of DCmotors. Also the embodiment of the invention can be applied to what isknown as universal motors, a motor with conventional brushes that isinsensitive to the sense of the current through it, and it is normallydriven by an AC voltage source.

In yet another modification, the capacitor (65) in FIG. 6 can bepresent, not as an individual component, but embedded in other circuitblocks, as for instance the electronic commutator of a brushless DCmotor if the motor (66) is a brushless DC motor, or other controlelectronics coupled with the motor (66).

All the strategies of pulse-width modulation used with the known systemscan be used with the embodiment of the invention to implement thecontrol system (68) and the pulse width modulator (69). Similarly, theswitch (62), the diode (63) and the current sensor (67) have the samefunction as with the current state of the art and can be designedfollowing criteria well known to those skilled in the art.

Some of the numerous possible variations of the circuit shown in FIG. 6are shown in FIG. 7A to FIG. 7L.

FIG. 7A shows a system similar to FIG. 6 with a different topology. Thepower source (7A01) has a similar function to the voltage source (61) inFIG. 6. Motor (7A02) has a similar function to the motor (66) in FIG. 6.Capacitor (7A03) has a similar function to the capacitor (65) in FIG. 6.Diode (7A04) has a similar function to the diode (63) in FIG. 6. Theinductive element (7A05) has a similar function to the inductance (64)in FIG. 6. Control system (7A07) has a similar function to the controlsystem (68) in FIG. 6. Pulse width modulator (7A06) has a similarfunction to the modulator (69) in FIG. 6. Current sensor (7A09) has asimilar function to the current sensor (67) of FIG. 6. The switch (7A08)has a similar function to the switch (62) of FIG. 6, however, switch(7A08) is connected to ground rather than the positive terminal of thepower source (7A01). This circuit makes the implementation of theelectronics switch (7A08) and its drivers easier (or simpler) than theelectronic switch (62) in FIG. 6. However, the motor (7A02) is removedfrom the common terminal of the circuit.

Although electrically it appears to be similar to swap the position ofthe inductive element (7A05) and the parallel combination of the motor(7A02) and the capacitor (7A03), in practice, it is better to connect asshown in FIG. 7A. The reason for this is that in the way shown, theisolation of control elements connected to circuits in the motor aresubject to less AC voltage stress with respect to grounded circuitblocks.

A detailed implementation of this topology is shown in FIGS. 9A to 9L.

FIG. 7B shows that there are many points to measure the current for thecontrol system of the circuit disclosed in FIG. 6. FIG. 7B follows thesame architecture as FIG. 6. In this figure, the blocks (68) and (69) ofFIG. 6 have been joined together in a single control system (7B09).

(7B01) is similar to (61). (7B10) is similar to (62). (7B06) is similarto (64). (7B11) is similar to (65). (7B12) is similar to (66). (7B7) issimilar to (63).

All the components 7B02, 7B03, 7B04, 7B05 and 7B06 are current sensors.However, as will be apparent to those skilled in the art, not all thepositions render the same waveform. Furthermore, if the sensor is notsensing current in a branch of the circuit connected to ground (e.g., in(7B02) and (7B06)), differential amplification techniques like thecircuit shown on (7B14) must be used. The current in (7B03), (7B05) and(7B04) can be sensed with a simple small value resistor withoutdifferential techniques. A current transformer like the one illustratedin (7B13) can be used in all the locations. Also, the current in (7B02)flows when the current in (7B03) is not flowing. Current in (7B06) issimilar to (7B04). With the values normally chosen for the capacitor(7B11), at the frequency of operation normally chosen to control theswitch (7B10), all the high frequency components of the current flowthrough (7B11) and the low frequency components flow through the sensorin (7B05).

Hence a sensor in (7B05) cannot be used to control the current on acycle-by-cycle basis, but only as the average over a certain number ofcycles of the pulse width modulator embedded in (7B09). Reference 10,chapters 13 and 14, pages 3.172 to 3.192 provide useful information forimplementation of current sensors.

FIG. 7C shows a system similar in topology to FIG. 6. In this system,the pulse width modulator (69) and the control system (68) of FIG. 6 areincluded in the control system 7C08. If it is compared with FIG. 6above: (7C01) is similar to (61); (7C02) is similar to (62); (7C04) issimilar to (64); (7C06) is similar to (65); (7C05) is similar to (66);and current sensor (7C07) is similar to (67).

The difference with FIG. 6 is a switch (7C03) in the position of thediode in the other figures (63, 7A04 and 7B07). Control system (7C08)has an extra output to control the additional switch (7C03). Thoseskilled in the art of designing switched mode power supplies willrecognize the technique known as “synchronous rectification”. Thistechnique consists of replacing the diodes by switches that dissipateless power than the diodes. The switches have to be closed or opened atthe precise moment that the diode that they replace conducts (switchcloses) or is blocking (switch opens). This technique is currentlypopular when the current is high, to increase the efficiency of switchedmode power supplies, especially at low output voltages. In practice, asthe power MOSFET transistors normally used to implement the switchesalso include a diode in parallel that can conduct when the switchperforming the synchronous rectification is not yet closed, thesynchronization does not need to be perfect. The control signals to theswitches are the logical complement of each other. The controller (7C08)must not close both switches (7C02) and (7C03) at the same time or thepower source (7C01) is shorted to ground. Each time the switches mustchange their state, a period of time in which both switches arecommanded to be open must be introduced. This time is what is normallyknown as “dead time”.

In the case of the switch replacing the diode (7C03), the diode inparallel with the switch starts conducting before the switch is made toclose. This causes a negligible loss in efficiency. To facilitate theimplementation, switch (7C03) can have an actual diode in parallel inaddition to the diode embedded in the MOSFET transistor.

Reference 12, page 60 gives details of synchronous rectification. Also,Reference 24, section 2.2.3, pages 2-76 refers to “An introduction tosynchronous rectifier circuits using PowerMOS transistors”, andReference 2 provides an implementation of synchronous rectification.

FIG. 7D shows a generalization of the circuit shown so far. The pulsewidth modulators (69) in FIG. 6 and (7A07) in FIG. 7 and the electronicswitch (62) or (7A08), for instance, are simply providing alternativelya voltage close VDC or an open circuit (close to 0 voltage, as measuredfrom the cathode of the diode (63) in FIG. 6).

Referencing FIG. 6, or FIG. 6B, any technique that provides acontrollable pulse width modulated waveform in the point (A), extractingpower from the DC power source (61) to be delivered to the componentsconnected to point A, can be used with results similar to what thesystem in FIG. 6 can achieve with the switch (62).

A magnetic device, like a transformer, coupled with a plurality ofswitches and diodes can be used to that end. In view of the foregoing,it is possible to generalize and define a Power Pulse Width Modulator(PPWM) block (7D02) comprising a plurality of switches in block (7D03)coupled with a magnetic device, also coupled to a plurality of diodes,the latter combination in block (7D04).

Blocks like (7D02) are currently found in switched mode power supplies.The switches in block (7D03) inside PWM (7D02) are controlled from thecontrol system 7D09.

The control system (7D09) includes the control system (68) and the pulsewidth modulator (69). It differs from similar blocks found in FIGS. 6and 7A, in that if there is more than one switch, the system has to havemore than one control line to command it to open or close.

Not shown in FIG. 7D, for sake of clarity, is the fact that in somecombinations with more than one switch, it may be necessary to measurethe current in more than one point. In those cases, control systems like(7D09) also differ from the combination (68) and (69) of FIG. 6, in thatthey receive current information from more than one sensor.

All other parts of the system shown in FIG. 7D are equivalent infunction to part of the system shown in FIG. 6. Specifically, withreference to FIG. 6: (7D01) is similar in functionality to the powersource (61); (7D06) is similar in functionality to the inductive element(64); (7D08) is similar in functionality to capacitor (65); (7D07) issimilar in functionality to motor (66); (7D10) is similar infunctionality to current sensor (67); and (7D05) is similar infunctionality to diode (63).

The concept behind FIG. 7D will be more clearly understood in light ofthe following figures showing topologies currently used with switchedmode power supplies applied to the circuit disclosed in FIG. 6. In eachcase, there is a different implementation of the PPWM block (7D02).

The reasons for adding a transformer to the circuit disclosed in FIG. 6are:

-   -   Possibility of isolation between the power source (7D01) and the        motor (7D07) or the control system (7D09).    -   The voltage of the power source (7D01) is not limited to values        larger than the operating voltage of the motor (7D07).

The reason for not simply using an off-the-shelf switched mode powersupply to drive the motor in current control mode using the circuitshown in FIG. 5 is that by implementing the circuit in the mannerdisclosed, a simpler, less expensive system results, especially if thesystem has to provide quick accelerations of the motor and power factorcorrection.

The current levels in the secondary side of the transformers (the motorside) are observed in the primary side (the switch or switches side)affected by the turns ratio of the transformer. This must be taken intoaccount while designing the switches, and when placing current sensorsin the primary side of the transformers.

After the generalization made in FIG. 7D, four examples follow:

Example 1 FIG. 7E. Topology Derived from the Forward Converter

The system in FIG. 7E uses a transformer (7E03) to couple a rectangularwaveform to the node labelled (A) in the drawing. There is a pulse widthmodulator embedded in the control system (7E11) that controls the switch(7E09) that can command it to open or close. When the switch (7E09)opens and closes, a rectangular waveform is applied to the primary sideof the transformer (7E03). This rectangular waveform is coupled by atransformer (7E03) to its secondary side and is coupled to the nodelabelled (A) in the drawing, through the diode (7E04). The node labelled(A) in the drawing is coupled to a system similar to the one found inFIG. 6. If FIG. 7E is compared with a combination of FIG. (6) and FIG.(7A), the following applies: Power source (7E01) is similar infunctionality to the power source (61); winding or inductive element(7E05) is similar in functionality to the inductive element (64);capacitor (7E07) is similar in functionality to capacitor (65). Motor(7E08) is similar in functionality to motor (66); (7E10) is similar infunctionality to current sensor (7A09); (7E09) is similar infunctionality to the switch (7A08); and (7E06) is similar infunctionality to diode (63). The pulse width modulator (7A06) and thecontrol system (7A07) are embedded in the control system (7E11).

The function of diode (7E04) is to prevent the winding of transformer(7E03) short-circuiting the DC voltage in the point labelled (A) in thedrawing to ground. Diode (7E02) discharges the energy accumulated in theinductance of the transformer (7E03), while the switch (7E09) is open.With this type of connection, the duty cycle of the pulse widthmodulator inside (7E11) has to be less than 0.5 (50%).

The same pulse width modulator detailed in FIG. (6B) can be used in(7E11), using the small resistor (67) of FIG. 6B to implement the sensor(7E10). However, it must be kept in mind that the current sensed in theposition of sensor (7E10) is the current though the inductive element(7E05), affected by the number turn relationship of transformer (7E03).

Clearly, the combination of (7E02), (7E03), (7E04) and (7E09) implementsthe block (7D02) of FIG. 7D.

Those skilled in the art of switched mode power supplies, rather thanmotor control, will recognize that the system of FIG. 7E has part of thetopology of a current mode “forward converter”.

Details of how to design each of the components coupled with thetransformer (7E03) and the transformer itself can be found in thefollowing references:

Reference 3, page 217.

Reference 7, pages 104 and 170.

Reference 9, page 37.

Reference 10, pages 2.63 to 2.69.

Reference 11, pages 76-83.

Reference 12, pages 32 and 40.

Reference 24, pages 2-13, 2-84 and 2-101.

Example 2 FIG. 7F. Topology Derived from the Push-Pull Converter

FIG. 7F uses a transformer (7F04), two switches (7F02) and (7F12), andtwo diodes (7F05) and (7F06) to implement the block (7D02) in FIG. 7D.The two switches alternate in opening and closing. The combination ofelements named above generates a pulse width modulated waveform in thenode labelled (A) in the drawing.

The node labelled (A) in the drawing is coupled to a system similar tothe one found in FIG. 6. If FIG. 7F is compared with a combination ofFIG. 6 and FIG. 7A, the following applies: (7F01) is similar infunctionality to the power source (61); (7F08) is similar infunctionality to the inductive element (64); and (7F09) is similar infunctionality to capacitor (65). (7F10) is similar in functionality tomotor (66); (7F13) and (7F03) are similar in functionality to currentsensor (7A09); (7F02) and (7F12) are similar in functionality to theswitch (7A08); and (7F07) is similar in functionality to diode (63). Thepulse width modulator (7A06) and the control system (7A07) are embeddedin the control system (7F11).

The function of diodes (7F05) and (7F06) is to prevent the winding oftransformer (7F04) from short-circuiting the DC voltage in the pointlabelled (A) in the drawing to ground.

The control system (7F11) has the functionality of the control system(7A07) and the pulse width modulator (7A06). One difference is thatcontrol system (7F11) has to take information from two current sensors(7F03) and (7F13). However, since when one switch is open the other isclosed, the current information from both sensors can simply be addedtogether, and can be filtered with a low pass filter, to get a waveformsimilar to the waveform of sensor (67) in FIG. 6. It must be kept inmind that the current sensed by the sensors is the current through theinductive element (7F08), affected by the number of turns relationshipof the transformer (7F04). The other difference is that consistentlywith the use of two switches, and as it was mentioned while commentingon FIG. 7D, the control system (7F11) has two outputs to control oneswitch each.

The system formed by transformer (7F04), diodes (7F05) and (7F06) andthe switches (7F02) and (7F12) can be analyzed and designed withtechniques similar to the techniques used in a “push-pull” switched modepower supply.

Details of how to design each of the components coupled with thetransformer (7F04) and the transformer itself can be found in thefollowing references:

Reference 3, page 220.

Reference 7, pages 116 and 153.

Reference 9, page 37.

Reference 10, pages 2.147 to 2.151 and 2.153 to 2.159.

Reference 11, pages 34-38.

Example 3 FIG. 7G. Topology Derived from Half- and Full Bridge Converter

In FIG. 7G, a block equivalent to the block (7D02) of FIG. 7D isimplemented with capacitors (7G02) and (7G14), switches (7G03) and(7G04), diodes (7G06) and (7G07) and transformer (7G05).

The two switches alternate in opening and closing. The combination ofelements named above generates a pulse width modulated waveform in thenode labelled (A) in the drawing.

The node labelled (A) in the drawing is coupled to a system similar tothe one found in FIG. 6. If FIG. 7G is compared with a combination ofFIG. 6 and FIG. 7A, the following applies: (7G01) is similar infunctionality to the power source (61); (7G09) is similar infunctionality to the inductive element (64); and (7G11) is similar infunctionality to capacitor (65). (7G10) is similar in functionality tomotor (66); (7G12) is similar in functionality to current sensor (67);(7G03) is similar in functionality to the switch (62) and (7G04) issimilar in functionality to the switch (7A08); and (7G08) is similar infunctionality to diode (63).

The pulse width modulator (69) and the control system (68) are embeddedin the control system (7G13).

The function of diodes (7G06) and (7G07) is to prevent the winding oftransformer (7G05) to short-circuit the DC voltage in the point labelled(A) in the drawing to ground.

The control system (7G13) has the functionality of the control system(68) and the pulse width modulator (69). The current sensor 7G12 can beimplemented with the small resistor of FIG. 6B. The control system mustcontrol two switches, so as mentioned while commenting on FIG. 7D, thereare two control outputs.

Those skilled in the art of switched mode power supplies, rather thanmotor control, will recognize the transformer arrangement of the“half-bridge” switched mode power supply. Furthermore, by looking atFIG. 7G, it will be obvious to those skilled in the art of switched modepower supplies, rather than motor control, that replacing the capacitors(7G03) and (7G14) with diodes with the cathode connected towards thepositive terminal of the power source (7G01) provides a “diagonal-halfbridge” (sometimes called a “two transistor forward converter”).Further, by replacing the capacitors (7G03) and (7G14) with electronicswitches controlled by adding two extra switch control lines to thecontroller in FIG. 7G13, the “bridge”—also known as a “full bridge”configuration—is obtained.

Details of how to design each of the components coupled with thetransformer (7G05), and the transformer itself, can be found in thefollowing references:

Reference 3, page 223.

Reference 7, pages 111, 113 and 152.

Reference 9, page 93.

Reference 10, pages 2.80 to 2.115.

Reference 12, page 33.

Example 4 FIG. 7H. Topology Derived from the Flyback Converter

This embodiment is based on the switched mode power supply topologyknown as a flyback converter (also known as “ringing choke powersupply”). In this case, the system is different from FIG. 7D, since asingle component, the transformer (7H03), provides the functionality ofboth, the inductive element (64) and the catch or freewheeling diode(63) in FIG. 6. The inductive element (64) is provided by thetransformer inductance, and the diode (63) is not needed because of theproperties of the magnetic system formed by the transformer (7H03) ifits windings are connected with opposing polarities. Transformersworking in which the mode (7H03) is working have also been described inthe literature as two coupled inductors.

Diode (7H04) is used simply to prevent the secondary winding of thetransformer to short-circuit the positive motor voltage to ground.

Snubber circuits in blocks (7H02) and (7H07) are needed due to the factthat a practical transformer (7H03) has parasitic components, like forinstance leakage inductance.

If FIG. 7H is compared with a combination of FIG. 6 and FIG. 7A, thefollowing applies: (7H01) is similar in functionality to the powersource (61); the functionality of the inductive element (64) is providedby transformer (7H03); (7H05) is similar in functionality to capacitor(65). (7H06) is similar in functionality to motor (66); (7H09) issimilar in functionality to current sensor (7A09); (7H08) is similar infunctionality to the switch (7A08); the functionality of diode (63) isprovided by transformer (7H03). The pulse width modulator (7A06) and thecontrol system (7A07) are embedded in the control system (7H110).

The control system (7H110) can be implemented with the pulse-widthmodulation method shown in FIG. 6B. This is a frequently used method forcontrolling current through a flyback transformer. An integrated circuitoften used to implement this functionality in switched mode powersupplies is the UC2845 (see Ref. 23).

Details of how a flyback transformer operates, how to design each of thecomponents coupled with the transformer (7H03), and the transformeritself can be found in the following references:

Reference 3, page 214.

Reference 7, pages 151, 166 and 190.

Reference 9, page 105.

Reference 10, pages 2.3 to 2.62.

Reference 11, pages 29-34 and 83-90.

Reference 12, pages 32, 36-34 and 83-90.

Example 5 FIG. 7J. New Topology Derived from an Un-Insulated InputReferenced Flyback Converter or from an Input Referenced Boost Converter

This embodiment is loosely based on the switched mode power supplytopology known as a flyback converter (also known as “ringing chokepower supply”). However, it is different in the sense that it does notuse a transformer and the output is referred to the input and not toground.

The circuit works as follows: When the switch (7J06) closes, currentflows from the DC voltage source (7J01) to the inductive element (7J05).Diode (7J04) is reverse-biased. When the switch (7J06) opens, thecurrent circulating through the inductive element flows through thediode (7J04), to the parallel combination of the capacitor (7J02) andthe motor (7J03).

If FIG. 7J is compared with FIG. 7A, the following applies: (7J01) issimilar in functionality to the power source (7A01); the functionalityof the inductive element (7A05) is provided by (7J05); (7J02) is similarin functionality to capacitor (7A03); (7J03) is similar in functionalityto motor (7A02); (7J07) is similar in functionality to current sensor(7A09); (7J06) is similar in functionality to the switch (7A08); thefunctionality of diode (7A04) is provided by (7J04). The pulse widthmodulator (7A06) and the control system (7A07) are embedded in thecontrol system (7J08).

The control system (7J08) can be implemented with the pulse-widthmodulation method shown in FIG. 6B.

There is currently no switch-mode power supply topology using aconfiguration from which the circuit can be derived. An analysis of thecircuit operation is provided in FIG. 15, for the case of continuouscurrent flow though the inductance (7J05).

Once the analysis is done, it can be learned from it that the dynamicsof the system can be described with equations similar to the buck-boostconverter (also known as the positive to negative converter). However,care should be taken to keep in mind the polarity of the output and thefact that it is not referenced to ground. The analysis of the dynamicbehavior of the buck-boost converter (also known as the positive tonegative converter) can be found in References 3, 9 and 10.

The meaning of the notation used in FIG. 15 can be seen directly fromthe circuit diagrams in FIG. 15. The analysis is made for the case ofcontinuous current circulation through the inductive element. The resultin equation (1506) shows that the voltage over the motor V_(M), can belower or higher than the value voltage source (7J01). The relationshipbetween both voltages depends on the relationship between T_(ON) andT_(OFF).

The Equations (1501) and (1502) are similar to the equations (3) and (4)in FIGS. 6C-1. Equation (1503) shows that it is assumed that the voltagedrop in the inductive element (7J05) is much larger in value than thevoltage drop in the switch (7J06) and the current sensor (7J07).Similarly equation (1504) shows that it is assumed that the voltage dropin the diode (7J04) is much smaller than the value of the voltage overthe motor.

It will be obvious to those skilled in the art how to do the sameanalysis for the discontinuous case, wherein the current through theinductive element drops to zero before T_(OFF) finish. This type ofanalysis is common to switch-mode power supply technology. In any case,the results from the analysis of the buck-boost converter (negative topositive converter) can be used with the precautions explained above.

Example 6 FIG. 7L. Topology Derived from the Buck-Boost Converter

This embodiment is based on the switched mode power supply topologyknown as a buck-boost converter (also known as a positive to negativeconverter). The circuit works as follows: When the switch (7L06) closes,current flows from the DC voltage source (7L01) to the inductive element(7L05). Diode (7L04) is reverse-biased. When the switch (7L06) opens,the current circulating through the inductive element will flow throughthe diode (7L04) to the parallel combination of the capacitor (7L02) andthe motor (7L03).

If FIG. 7L is compared with a combination of FIG. 6 and FIG. 7A, thefollowing applies: (7L01) is similar in functionality to the powersource (61); the functionality of the inductive element (64) is providedby (7L05); (7L02) is similar in functionality to capacitor (65). (7L03)is similar in functionality to motor (66); (7L07) is similar infunctionality to current sensor (7A09); (7L06) is similar infunctionality to the switch (62); the functionality of diode (63) isprovided by (7L04). The pulse width modulator (69) and the controlsystem (68) are embedded in the control system (7L08).

The control system (7L08) can be implemented with the pulse-widthmodulation method shown in FIG. 6B.

For this embodiment there is a well-known switch-mode power supplytopology from which the steady state operation can be derived (see Ref.9, page 150, Ref. 10, page 31 and Ref. 3, page 81).

For the case of continuous current through the inductance, the resultsare similar to the circuit 7J. Hence the voltage over the motor can belarger or smaller than the voltage of the DC power source.

As with all the other examples, the voltage over the motor for the givencurrent set by the control system (7L08) will depend on thecharacteristics of the motor, its load, and the speed of rotation.

The advantage of this circuit with respect to the circuit in FIG. 7J isthat the motor is referred to ground, however, the voltage is ofnegative polarity with respect to the voltage of the DC power source.

FIG. 7I illustrates a different embodiment of the circuit disclosed inFIG. 6. The main difference is that the current through the motor (7I7)as measured with the current sensor (7I12) is controlled independentlyof the current through the inductance L_(M) (7I5), which is measuredwith the current sensor (7I11). The current through the inductance (7I5)is controlled by applying a pulse width modulated waveform to thecontrol terminal of the switch (7I3) in a way similar to FIG. 6. Thecurrent through the motor is controlled by applying pulse widthmodulation techniques to the switches in the electronic commutator (7I6)in FIG. 7I. In this case, the motor current is controlled using theinductance of the motor windings, exactly as the current state of theart does. This is shown in FIG. 5. FIG. 7I shows two different controlsystems, for sake of clarity. However, in practice, a singlemicroprocessor can control both current loops. In this case, the setcurrent signal shown from the main controller (7I10) to the controller(7I8) is a value passed between different segments of a running program.

The system of FIG. 7I can revert to the system of FIG. 7A, if theelectronic commutator (7I6) of the motor (7I7) is not pulse widthmodulated. In this case the current through the motor 7I7 is the lowfrequency component of the current switched through the inductance L_(M)(7I5) by the switch (7I3) and the diode (7I4).

If the current through the motor (7I7) is made different than thecurrent provided through the coil (7I5), charge accumulates in capacitorC_(M) (7I9) raising the voltage of the capacitor.

Hence the current through the motor can still be controlledindependently of the current through the coil L_(M), in all the otherembodiments, without sensing the actual motor current, simply bymeasuring the voltage in the capacitor with relation to what the voltageshould be if it were the product of the back electromotive force inducedvoltage in the motor windings.

Allowing control of the current through the inductance (7I5)independently of the current provided to the system comprising the motor(7I7), its electronic commutator (7I6) and the capacitor (7I9) yield avery versatile apparatus. There are at least two applications for such asystem:

Application_7I_1. Power Factor Correction

The capacitor in the rectifier filter (7I2) is chosen small enough forthe voltage at the output of the rectifier (7I1) to track the full wave,rectified AC main voltage. The current in the inductance L_(M) (7I5) ismade a full wave rectified waveform:I(t)=Ipeak*|sin(φ)|where the current waveform is made synchronous with the full waverectified AC line.

The controller (7I10) has to make the mean value of the current waveform(Ipeak* 2/π) equal to the average current that is taken by the motor(7I13) and control electronics, from the capacitor C_(M) (7I9).

The current state of the art uses an analog multiplier to multiply theattenuated full wave AC rectified waveform by the set current for thecontrol loop (see Refs. 20 and 21).

A proposed method is to sense the time of zero crossing of the AC signaland calculate the phase (φ) of the sinusoidal signal by relating thetime at which a new current value must be set with the period of theline. The period of the line can be measured continually calculating thetime it takes for the number of zero crossing events to occur anddividing the measured time by that number (e.g., 16 is a good choice forthat number).I(t)=I_average_needed*π/2*|sin(ticks_from_(—)0_crossing/ticks_in_period)

Where:

-   -   I_average_needed is the average current required through the DC        motor; ticks_from_(—)0_crossing is the amount of time from the        zero crossing of the AC main measured in an arbitrary unit        (events of a periodic phenomena or “ticks”);    -   ticks_in_period is the period of the AC main measured in an        arbitrary unit (events of a periodic phenomena or “ticks”).

In this way there is no need for connecting an analog to digitalconverter input of the controlling microprocessor to the AC main.

Table 1 contains a pseudo-code, following the syntax of the C language,showing the algorithm.

TABLE 1 // Power Factor Correction Algorithm without A/D converterInitialization( ) { P_Ticks = 0 ; // clear ticks accumulator Ticks = 0 ;// clear ticks accumulator Period = 0 ; // set initial value of periodas not calculated yet Old_Line = 0 ; // set initial value for algorithmN_cycles = 0 ; // clear 16 cycle counter Set timer interrupt( ) ; //initialise the hardware for a periodic timer interrupt } // End ofinitialization Timer_interrupt( ) { // here each timer interrupt ;P_Ticks = P_Ticks + 1 ; // count for calculating the period Ticks =Tick + 1 ; // count for calculating the phase New_Line = Read_line_state( ) ; // read state of the AC line − high or low If ( New_line <>Old_line ) {   // here if there was a change in state of the AC main  Old_line = New_line ; // save state for detecting next transition   If( New_line = High ) {     // here if a new AC line cycle     Ticks = 0 ;// clear the phase accumulator     N_cycles = N_cycles + 1 ; // countthe line cycles     If ( N_cycles > 15 ) {     // here after 16 linecycles     // a new period average can be calculated     N_cycles = 0 ;// clear cycle counter     Period = P_Ticks / 16 ; // calculate theperiod average of last 16 cycles     P_Ticks = 0 ; // clear accumulatorof next calculation           }         } If ( Period > 0 ) { // here tomodulate the current      I_out = I * π / 2 * | sin ( Ticks / Period ) |;     }   else     { // here the first 16 cycles, no PFC      I_out = I;     } // Note: “I” is calculated elsewhere in a control algorithmsuitable for the application } // end of the timer interruptApplication_7I_2. Bilevel CPAP Device

In a bilevel device, the motor needs to take considerable current whileaccelerating to raise the air pressure delivered to the patient. After afew seconds, the motor is braked for decelerating to drop the airpressure. While the current through the motor varies considerably, theaverage current through the inductance needs only to be equal to theaverage current drawn by the motor and the control system (normally thecurrent drawn by the control system is negligible in comparison with thecurrent drawn by the motor).

This application can also make use of the power factor correction methodexplained above.

Application_7K. Power Factor Corrected Brushless DC Motor Controller

The system in FIG. 7K makes use of the topology of FIG. 7J and thetechnology disclosed in FIG. 7I. The DC voltage source (7J01) in FIG. 7Jis equivalent to the output of the full wave rectifier (7K01) in FIG.7K. The capacitance of capacitor Ci (7K02) is of small value so thevoltage in the node X tracks the full wave rectified AC mains waveform.

The functionality of the switch (7K07) is similar to (7J06) and also tothe switch (7I3) in FIG. 7I. However, the switch (7K07), like the switch(7J06), is a low-side drive switch, similar to the switch (7A08). Thecurrent sensor (7K08) has a similar functionality to the sensors (7I11),(7J07) and (7A09).

The inductive element (7K06) has similar functionality to (7J05). Thecapacitor (7K05) has a similar functionality to the capacitors (7I9) and(7J02).

The feedback circuit block (7K09) can be implemented by using a pulsewidth modulated output from a microcontroller and linking it through anoptocoupler with the set point input of the power factor correctioncontrol (7K03). A practical example of this type of link can be found inFIG. 9H.

The function of the controller (7K03) is similar to the function of thecontroller (7I10) combined with (7I8), as described for FIG. 7I.

The controller (7K03) shapes the current through the inductive element(7K06) so the system has a power factor close to unity.

Any of the current state of the art techniques or algorithms for powerfactor correction can be used in the controller (7K03) (see Ref. 10,page 222 or Ref. 25, chapter 1):

-   -   Fix-on time, Discontinuous Current Control (DCC), with or        without fixed output voltage (for the boost topology is called a        “boost follower”).    -   Critical Conduction Mode (CRM) also known as Transitional Mode        Controllers (Ref. 25, page 8).    -   Continuous Conduction Mode (CCM) Control (see Refs. 20, 21 and        25).

In continuous current control, the control loop can use as feedbackseveral alternatives, for instance average current control or peakcurrent control.

From a practical implementation point of view, most if not all thecurrent integrated circuits controllers designed for boost converterbased power factor correction circuits can be used with littlemodification for the block (7K03).

If the power factor correction controller is based on a microcomputerwith analog to digital converter the technique disclosed in thedescription of the controllers of FIGS. 7I and 9 can be used.Alternatively, if the factor correction controller is based on amicrocomputer without analog to digital converter the algorithmdisclosed in TABLE 1 can be used.

If a classical controller with fix oscillation frequency, set byexternal components, is used along with a microcomputer, the techniqueshown in FIGS. 9H, 9I and 17F, in combination with the algorithm ofTABLE 3, can be used to change randomly the frequency of oscillation ofthe controller, improving on the EMI characteristics of the system.

The block (7K10) is a brushless DC motor controller. (7K12) is thebrushless DC motor. A current sensor (7K11) for the current through themotor (7K12) is shown in FIG. 7K. However, a similar system would workif no direct measurement is made of the current though the motor (7K12).In this case, the current would be assumed by design and adjusted by themain controller (7K14) in response to another measured physical quantityobtained through the sensors (7K15). If there is no need for fastreaction from the brushless DC motor (7K12), the system can be furthersimplified by removing the pulse width modulation from the controller(7K10). That is a slow reacting system that still works if no pulsewidth modulation is applied to the windings L1 to L3 of the motor(7K12). In this last case, the current through the motor is controlledby controlling the average current through the inductance (7K06). Thecurrent through (7K06) tracks the full wave rectified pulsating voltageat the node X. The inertia of the motor plus its load filters thecurrent pulses. This last case is ideal for cooling or ventilation fans,while the full control scenario (usage of sensor (7K11) and pulse widthmodulation of the winding in (7K12)) is ideal for bi-level flowgenerators for the medical industry.

Capacitor (7K13) is included to show that since the controllers (7K14)and (7K10) and the motor (7K12) itself are not connected directly toground, a plurality of de-coupling capacitors with good high frequencyresponse, connected to a ground plane, may be necessary to improve theability of the set-up to pass the electromagnetic compatibilityregulations.

FIGS. 8A to 8C show the relationship between the circuit and thetechnology used in switched mode power supplies (SMPS), as applied toflow generators. To facilitate the comparison, in the three drawings ofFIG. 8, blocks with the same number have a similar function.

FIG. 8A shows a flow generator with a switched mode power supply (SMPS)controlled in voltage mode. The controller 8A1, in the SMPS, usesfeedback from the output voltage of the SMPS (Vout) to control a pulsewidth modulator (PWM). The PWM (85) operates a plurality of switches(82) that interact with a magnetic system (83). The controller (88)controls the current through the BLDCM (86) by applying pulse widthmodulation (8A2) to the electronic commutator (89), in response tofeedback obtained from a physical variable (87).

In FIG. 8B, there is a similar system with the exception of the SMPS.The only change is in the SMPS itself. In fact, the systems in FIGS. 8Aand 8B could interchange the SMPSs provided that both SMPSs meet thesame minimum requirements.

In the current mode SMPS, the feedback from the output voltage is usedby a controller (8B1), but the output of the controller yields the setcurrent (or desired current) of a current mode controller (8B5). Thiscurrent mode controller may use some common circuitry with someembodiments of the circuit, for instance FIGS. 7E, 7F and 7H. Currentmode control is a superior choice and its advantages for SMPS are wellknown.

The circuit applied to a CPAP flow generator is shown in FIG. 8C. Thereare several blocks in common, but unless a more flexible system like theone already disclosed in FIG. 7I is implemented, the followingdifferences can be seen:

D8_1 The voltage over the motor (VM) does not need to be a controlledvariable. The voltage is the consequence of the speed of rotation. D8_2The PWM in the electronics commutator (8B2 and 8A2) is not required.D8_3 The control system of the power supply itself is fully integratedinto the main controller (88).

Once the flow generation is in operation, a voltage can be measuredacross the motor. Since the control electronics need only a smallfraction (say, as a example, around 5%) of the power needed to operatethe motor, the voltage measured across the motor, as the result of itscontrol operation, can be used to extract the power necessary forcontrol electronics. This is the concept disclosed in FIG. 4 anddetailed further in FIG. 10.

To power the control electronics (44 in FIG. 4 and 10A07 in FIG. 10A)while there is no voltage produced over the motor an “auxiliary powersupply” (42 in FIGS. 4 and 10A09 in FIG. 10A) is used. The idea is touse the “auxiliary power supply” to only provide power during theinitial phase of equipment operation after power is applied.

Once a suitable voltage on the motor is established, an electronicswitch (43 in FIGS. 4 and 10A08 in FIG. 10A) is used to unload the“auxiliary power supply”. Alternatively, the “auxiliary power supply”can be shut down by the control electronics itself (10A07) orautomatically (see circuit of FIG. 10B). If the power drawn by the“auxiliary power supply” is small, and it is acceptable for it to remainoperating, a simple current summing junction can replace the electronicswitch (10A08). The advantage of this scheme is that the “auxiliarypower supply” can be made simply and inexpensively, since the auxiliarypower supply operates or draws significant power only for a short periodof time. As the auxiliary power supply draws power during a short periodof time, the heat generated inside the case is small.

There is another form of operation for the system comprised by theelectronic switch and the auxiliary power supply. This system is shownin FIG. 10D.

In FIG. 10D, the auxiliary power supply has a capacitor (10D03) that ischarged with current from the AC main through a resistor (10D01) and arectifier (10D02). The switch (10D05) is normally open. A voltagecomparator with hysteresis (10D10) senses the voltage in the capacitor.The comparator (10D10) closes the switch (10D05) when the voltage in thecapacitor is larger than a certain value. When the switch is closed, theelectronics (10D12) is powered. Once the electronics are powered, thevoltage in the capacitor (10D03) decreases due to the power consumptionof the electronics. The electronics (10D12) have to activate the motordriver to extract power from the AC main and generate enough voltageover the motor, so a voltage adapter (10D13) can provide power to theelectronics (10D12) via the diode 10D04. The voltage adapter (10D13) istypically a simple linear voltage regulator but can be as complex as aDC/DC converter, depending on the fluctuations and the value of theavailable voltage.

The current state of the art would also use a capacitor like (10D03).Charged with a similar system, however, rather than a switch like(10D05), the electronics would be made to operate in a very low powermode. The advantages of this aspect of the circuit are:

SW_A1. There is no need for low power electronics in the circuitfollowing the switch. SW_A2. The switch (10D05) and the comparisonsystem (10D06) can be implemented in a simple way, with commonlyavailable discrete components at a lower cost and drawing a lot lesspower than a system using low power electronics. Low power electronicsneed to draw power to bias internal circuitry. The differences can bemicro-amps, for the low power electronics (due to the needs of currentbiasing circuits), in comparison with nano-amps for the switch (theleakage current of a semiconductor). It has to be noted, that in thecurrent state of the art of low power micro-controllers, the lower poweris drawn, only after normal power is drawn, to initialise the low powermode. Such devices only work with the switch option. The resistor(10D01) charging the capacitor (10D03) is typically never disconnected.The less power the system can draw from the capacitor, before operation,the larger the resistor value and the lower the power that iscontinually dissipated in (10D05). With the possible practicalimplementations of (10D05) and (10D06), the main problem is the leakagecurrent of the capacitor (10D03) itself. SW_A3. The currently availablelow power electronic devices also work from lower voltages than thehigher power consumption options. A practical switch can be implementedwithout that restriction and switch when a higher voltage is present inthe capacitor (10D03). That higher voltage in the capacitor allows theelectronics to operate longer before the other source of power isavailable.Advantages for an Airflow Generator:

The practical advantages of the embodiments of the invention for anairflow generator for CPAP are:

-   (1) Allow a simpler, less expensive power supply as the requirements    of the brushless DC motor (BLDCM) and its electronic commutator are    far less demanding than those of the controlling electronics. This    is particularly useful in bi-level devices as the sudden    acceleration of the motor requires extra power, and power supplies    tend to be bigger, more complex and expensive.-   (2) Allow the controlling elements of the power supply to be fully    integrated into the controlling electronics (typically    microprocessor based), hence lowering the component count, which in    turn leads to more reliability and lower cost.-   (3) Allow greater flexibility, since the controlling processor can    be made to control not only the current though the BLDCM, but also    the voltage over the BLDCM that can be made to change continually as    commanded by the control electronics. If as it is commonly used to    control the voltages over the windings, the electronic commutation    is pulse width modulated, it allows a flexible control scheme in    which simultaneously the voltage available to the pulse width    modulating circuit can be changed in addition to the pulse width    itself.-   (4) If the flexibility discussed above is not necessary, then the    electronic commutation of the BLDCM does not need to include pulse    width modulation. The resulting system is simpler, has fewer    components and is more efficient, since there is no power    dissipation in the switches (typically MOSFETs) of the electronic    commutator of the BLCDM when performing pulse width modulation.    Different Implementations:    I1. A Complete Flow Generator without Auxiliary Power Supply

FIG. 9 shows an example of an airflow generator for CPAP treatment thatuses the invention disclosed above. A conventional low power AC mainfrequency transformer with two secondary windings is used (9B2, in FIG.9B) to power the control electronics, while the BLDCM (9D2, in FIG. 9D)is driven from the rectified AC main (node VCCLINE, in FIG. 9A),following the circuit disclosed (see FIG. 9C) with the topology shown inFIG. 7A. The BLDCM (9D2, in FIG. 9D) Hall effect sensors and itselectronic commutator are powered from the voltage generated fromrunning the motor itself.

FIG. 9A shows the circuit from the AC main IEC power connector (9A6) tothe node VCCLINE. VCCLINE, the output of the circuit, is the full waverectified AC main. This is the DC voltage source (7A01) of FIG. 7A.

Two fuses (9A8) and (9A7) are included for protection. The switch (9A9)is used to turn the equipment on or off. The switched AC main voltage isfound between SA and SN. A metal-oxide varistor (MOV), (9A10) protectsthe device against AC main voltage transients.

A filter (9A1) is used for EMI compliance (Ref. 8). The requirements onthis filter should be reduced, as the frequency of the main source ofEMI is modulated using spread spectrum techniques. A rectifier bridge(9A2) connects as a full wave rectifier, with a capacitive filterprovided by (9A4). The capacitor (9A3) improves the high frequencyimpedance of the filter. A NTC (9A5) protects the rectifier bridge (9A2)against in-rush current when the equipment is turned on.

FIG. 9B shows the low power transformer (9B2) that provides power to thecontrol electronics. The nodes FMA and FMN in FIG. 9B are connected tothe input of the rectifier bridge (9A2) after the filter (9A1) and theNTC (9A5). A snubber network 9B1 protects the transformer. One of thesecondary windings is rectified by the bridge (9B3), and the output isfiltered by (9B6) and regulated by linear regulators (9B4) and (9B5).This circuit provides +15 volts to power the analog sensors, and +5volts to power the microcontroller. The sensors and the microcontrollerare in FIG. 9J.

FIG. 9C shows the other main components of the circuit. The coil (9C3)is the inductance L_(M), shown as (7A05) in FIG. 7A. The capacitor(7A03) of FIG. 7A is implemented with the capacitors (9C2) and (9C1).Capacitor (9C1) improves the high frequency behavior of the capacitor(9C2). Diode (9C4) corresponds to diode (7A04) of FIG. 7A. The motor(9C5) corresponds to the motor (7A02) of FIG. 7A.

Electrically, the BLDCM and the electronic commutator (also known asBLDCM “controller”) is a two terminal network with terminals labelledMOTOR_TOP and MOTOR_BT. The current flowing from MOTOR_TOP to MOTOR_BTis the variable used by the speed control loop, to control the speed ofthe BLCDM and hence the air pressure that the flow generator produces.

The switch of FIG. 7A is implemented by the power MOSFET (9H4) in FIG.9H. The current sensor (7A09) in FIG. 7A is implemented with theresistor (9H5) in FIG. 9H. (7A06) and (7A07) are implemented inside theUC2845 integrated circuit (9H3) in FIG. 9H. The data sheet of the devicemanufactured by Texas Instruments (it used to be a device manufacturedby Unitrode) can be consulted for details. The drain of (9H4) is thenode SW in FIG. 9H that connects to the same node in FIG. 9C. The“desired current” terminal in FIG. 7A is the command information thatthe microprocessor in FIG. 9J sends to the UC2845 in FIG. 9H.

FIG. 9D shows the interface with the BLDCM (9D2). A low voltage droplinear voltage regulator (9D1) gets +5 volts from the terminalMOTOR_TOP. The node is called (+5VSENSORS) in FIG. 9D. A diode (9D27),and a filter consisting of resistor (9D29) and capacitors (9D28), isused with the voltage regulator.

The motor sensors (terminals S1M to S3M of 9D2) interface the mainmicrocontroller through optocouplers (9D8, 9D9 and 9D10). The outputs ofthe optocouplers are the signals S1, S2 and S3 that connect to themicroprocessor (9J2) in FIG. 9J.

Part of FIG. 9D and FIGS. 9E and 9F show an implementation of anelectronic commutator for a typical BLDCM used in airflow generators forCPAP. The circuit shown is particularly flexible in terms of maximum andminimum voltage that the commutator would work with. This is achieved bythe use of a current source to drive the gate of the P-channel MOSFETs.

In FIG. 9D, resistor (9D25), Zener diode (9D26), resistor (9D24) andtransistor (9D23) form a current source. When the optocoupler (9D5) isnot conducting, the current in the collector of (9D25) causes Zenerdiode (9D11) to conduct, turning the P-channel MOSFET (9D3) on. Resistor(9D12) is included for protection.

The N-channel MOSFET (9D4) is of logic level type and is turn-on whenoptocoupler (9D6) conducts. A simple resistor (9D13) is shown in thegate of (9D4). However, extra circuitry to accelerate the removal ofcharge from the gate capacitance may be necessary if the pulse widthmodulation of the electronic commutator is used. The point connectingthe drain terminal of both power MOSFETs is connected to L1 terminal ofthe BLDCM; the signal name is L1C.

Negative voltage with respect to MOTOR_BT is used for the currentsource. The voltage is labelled MOTOR_BT_V and is generated from+5VSENSORS by the circuit shown in FIG. 9G. FIG. 9G shows a positive tonegative voltage converter using a cascade of two industry standardcharge pump integrated circuits ICL7660 (9G1 and 9G2 in FIG. 9G).

In FIGS. 9F and 9E, the current source, the N and P channel powerMOSFETs and the two optocouplers have a function similar to the circuitin FIG. 9E that drives the L1 winding of the motor. The L2 winding ofthe motor is driven by circuitry in FIG. 9E, while the L3 winding of themotor is driven by circuitry in FIG. 9F.

FIG. 9H shows a practical circuit that controls the switch (7A08) inFIG. 7A. An integrated circuit normally found in off-the-main switchedmode power supplies, the Unitrode's UC2845 (9 h 3), is controlling apower MOSFET (9H4). The current sensor (7A09) of FIG. 7A is the resistor(9H5). RC network (9 h 7) and (9 h 8) is a low pass filter that isnecessary to eliminate switching noise from the voltage resulting fromsensing the current.

The voltage of one of the windings of the transformer (9B2) is rectifiedby (9 h 1), filtered by (9 h 12) and regulated to 12 volts by (9 h 2).The 12 volts are used to power the current mode controller chip (9 h 3).

Transistor (9 h 12) saturates due to the current through resistor (9 h9), connected to the +12 volt supply. A low voltage in pin 1 of (9 h 3)inhibits the operation of the chip and keeps the power MOSFET (9 h 4)off. To start the operation of the chip, or to make it work in bursts,to achieve very low currents through the motor, the main microprocessoroperates the optocoupler (9 h 10) through resistor (9 h 11). A low levelin the terminal labelled (shdwn) enables the operation of the currentmode controller. In this way, the initial current can be set, before thechip can operate.

The current is set through pulse width modulation applied to resistor (9h 20), connected to the optocoupler (9 h 19). The waveform in thecollector of (9 h 19) is filtered by resistor (9 h 16) and capacitor (9h 17) and is buffered by operational amplifier (9 h 15) to give avoltage that is used by the error amplifier of the UC2845 (9 h 3). Theerror amplifier is programmed with a gain of (−1) by resistors (9 h 14)and (9 h 13). Capacitor (9 h 21) limits the bandwidth of the amplifier.The voltage in pin 1 of the UC2845 is reduced internally by subtractingtwo diode drops and divided by three before being compared by thevoltage sensed by the pin labelled (CS). When the voltage in (CS)exceeds the voltage in pin 1, as processed above, the switch (9 h 4) isturned off. The switch (9 h 4) is turned on by the oscillator inside theUC2845.

The optocouplers (9 h 10) and (9 h 19) are controlled by themicrocontroller (9J2) that controls the main function of the airflowgenerator. By changing the parameters of the pulse width modulation inthe optocoupler (9 h 19), the main microcontroller changes the maximumcurrent through the sensing resistor (9 h 5), hence changing the averagecurrent through the BLDCM. The other optocoupler (9 h 10) can be used toset the current to “0” for several cycles, hence allowing for smallaverage current to flow through the BLCDM. This optocoupler can also beused in case of error or when the equipment is in stand-by.

The timing for the oscillator is provided by an RC network connected tothe pin labelled (RC) in (9 h 3). Normally a simple RC network isconnected there.

FIG. 9I shows a circuit to change the frequency of operation of theUC2845, under the control of the microprocessor (9J2). This is done touse spread spectrum techniques to facilitate EMI compliance (Ref. 8).Pulse width modulation is applied through the optocoupler (912). Thewaveform in the emitter of the output transistor of (912) is filtered by(914) and (916). (913) and (915) form a resistive divider that limitsthe maximum voltage over the filter. The output is referred to +Vref,the 5 volts reference in the UC2845. Operational amplifier (917) andtransistor (9112) are connected as a current source that is proportionalto the voltage between +Vref and the output of the filter (voltage overthe capacitor (916)). This current charges the capacitor (9113) that ispart of the oscillator of UC2845. Resistor (9110) provides a minimumcurrent for charging the capacitor, giving a minimum frequency to theoscillator.

Another way of changing the frequency of the oscillator is to replacethe resistor in the RC oscillator of the UC2845 with a digitallycontrolled potentiometer.

FIG. 9-J shows the main microcontroller of the flow generator. Themicrocontroller (9J2) operates the electronic commutator of the BLDCM(9D2) and the user interface (9J3) and reads the information fromsensors (9J4). The circuit block (9J4) represents the pressure andairflow sensors typically found in the top end CPAP flow generators.(9J2) also interfaces with the current mode controller UC2845 of FIG.9H. This microcontroller could also communicate with other clinicalequipment (typically through serial lines) as is well-known practice.

The microcontroller (9J2) sets the current through the BLDCM by usingthe pulse width modulator PWM2 and an output line (signal name SETC andshdwn) connecting to the circuit of the UC2845. (9J2) also changes thefrequency of operation of the UC2845 by using another pulse widthmodulator (PWM3). The microcontroller (9J2) in FIG. 9J has to generatethe needed dead time, that is, the time necessary to prevent bothMOSFETs, the N-channel and the P-channel from conducting at the sametime, effectively short-circuiting MOTOR_TOP with MOTOR_BT.

A voltage to frequency converter, appearing as a block (9J6) in FIG. 9J,connects to microcontroller (9J2) through optocoupler (9J7). This isused to measure the voltage between MOTOR_TOP and MOTOR_BT.

FIG. 9K shows a circuit for the voltage to frequency converter (9J6).The voltage in MOTOR_TOP is filtered and regulated by a low voltage drop5V linear regulator (9K1). Operational amplifiers (9K16) and (9K22),along with transistors (9K21) and (9K6), form a controlled currentmirror. The current that charges the timing capacitor (9K8) depends onthe voltage across resistor (9K15), which in turn depends on the inputvoltage, the voltage between MOTOR_TOP and MOTOR_BT. The capacitor (9 k8) is connected to a CMOS 555 timer IC, (9K9). The capacitor (9 k 8) isdischarged by resistor (9K7), which is operated by the timer IC.Discharge time is constant and the charge time depends on the inputvoltage. Optocoupler (9K11) in FIG. 9K corresponds to optocoupler (9J7)in FIG. 9J.

A crowbar circuit (11A06), as shown in FIG. 11A, may be added, connectedbetween MOTOR_TOP and MOTOR_BT as a protection device. The crowbar is awell-known circuit in which an SCR is triggered when the voltage acrossthe terminals of the circuit exceeds a given value, working as aprotection against malfunctions. Usually when a crowbar is triggered, afuse (11A04 in FIG. 11A) is blown to prevent further damage.

An alternative way of operating the electronic commutator of the BLDCMis to use a “small”, 20-pin microcontroller (for instance, the AT90S2313from Atmel Semiconductor) to perform the logic and interface with themain controller via an optically coupled serial NRZ full duplex line.This solution releases the main microprocessor from the task ofoperating the electronic commutator. The 20-pin microcontroller canreport data to the main controller, such as speed of the BLDCM, as itwould be connected to the motor sensors. Commands to brake the motor oroperate coasting can be given by the main microcontroller to the 20-pinmicrocontroller through the serial communication line. Also withadditional interface circuitry, the “small” 20-pin microcontroller canreport the voltage over the motor or can operate its own pulse widthmodulator or current control circuitry, to have a system as wasdisclosed in FIG. 7I. The second “small” microcontroller can also beused as a complex watch dog timer. Both microcontrollers can reset eachother in case of detection of a fatal error condition.

FIG. 9L shows a low voltage drop linear regulator suitable to be used in(9D1), (9K1) and (11A17). The circuit is designed to be used with 24volt motors, since currently available integrated circuits such as, forinstance, the LM2940 can work continually only with 20 VCC.

In the circuit of FIG. 9L, an operational amplifier (9L21) works as theerror amplifier of a linear voltage regulator.

Resistors (9L26) and (9L27) form the feedback network, and itsrelationship determines the output voltage. (9L28) is a voltagereference that is powered from the output of the circuit itself throughresistor (9L24). A PNP topology is chosen for the pass transistor of thelinear regulator (9L5) to obtain low voltage drop. Transistor (9L7) andresistor (9L6) provide current limiting. The output current circulatesthrough resistor (9L6) and when the voltage drop reaches a value thatcauses the base-emitter junction of (9L7) to conduct, the currentthrough the collector of (9L7) diverts current from the base of (9L5),limiting the output current.

Resistors (9L9) and (9L8) bias transistor (9L5) from the drain terminalof a logic level MOSFET (9L13). The gate terminal of the logic levelMOSFET (9L13) is connected to a voltage source implemented with 3.3 voltZener diode (9L11), through resistor (9L12). Resistor (9L10) providesbias current to the 3.3 volt Zener diode (9L11). The collector currentof transistor (9L14) can cause a voltage drop in resistor (9L12) andturn off the logic level MOSFET (9L13).

On powering up the circuit, when the output voltage is lower than theminimum voltage needed for the operation of the operational amplifier(9L21) and transistor (9L17), the logic level MOSFET (9L13) is saturatedas its gate has the full 3.3 volts of the Zener diode (9L11). In thiscondition, maximum current circulates through the resistor (9L8) fromthe base of transistor (9L5). The current charges output capacitors(9L22) and (9L23). This arrangement guarantees that the circuit powersup. When the operational amplifier senses that the voltage at theoutput, as reduced by the resistive divider (9L26) and (9L27), is largerthan the voltage reference (9L28), the voltage on its output is raised.

The output of the operational amplifier (9L21) is connected to aresistive divider formed by (9L18), (9L17) and (9L16). The resistivedivider formed by (9L18), (9L17) and (9L16) biases the base oftransistor (9L17). Because of the emitter resistance (9L15), transistor(9L17) works as a current source. The higher the voltage differencebetween the sampled circuit output voltage in the non-inverting (+)input of the operational amplifier (9L21) and the reference voltage(9L28), connected to the inverting input of (9L21), the higher thevoltage at the base of (9L14) and the higher the current through itscollector.

The higher the current through collector of (9L14), the lower thevoltage in the gate of logic level MOSFET (9L13), and the lower thecurrent through the base of (9L5). The lower the current through thebase of (9L5), the lower the current through its collector, and thelower the output voltage of the circuit is, compensating the originaleffect. Capacitor (9L19) and resistor (9L20) compensate the feedbackloop, by shaping its frequency response. Resistor (9L1) and capacitors(9L4) and (9L3) form a low pass filter for the noise and voltagevariations of the input voltage V_(M). Schottky diode (9L2) is needed ifthe voltage available from V_(M) is discontinuous.

I2. Auxiliary Power Supply DC-DC Converter and Switch

FIG. 10B shows an example of the “auxiliary power supply” referred to inFIG. 10A and FIG. 4. This power supply typically dissipates significantpower during the first seconds of operation of the unit. It consists ofthe shunt regulator (Z2), formed by a programmable Zener diode TL431,that is followed by a power MOSFET (MF1) connected to the voltagegenerated by the TL431 via the resistor (R3). The combination of Zenerdiode (Z1), transistor (T1) and the resistors (R6) and (R7) causes thevoltage at the output to stabilize around 12 volts.

If the voltage at the output (point labelled APS) grows, the basecurrent of transistor (T1) also grows, causing the collector current oftransistor (T1) to grow. This, in turn, causes the voltage in the gateof MOSFET (MF1) to decrease, causing the output voltage to decrease. Inthis way, the system is stable. The electronic switch shown in FIG. 4 ismade with diodes (D1) and (D2) for sake of clarity.

When the linear regulator 7815 produces 15 volts, diode (D2) conductsand diode (D1) is reverse-biased. However, in a practical circuit, diode(D1) can be replaced by a short-circuit, since a voltage larger thanaround 13 volts in the output (node labelled APS) makes the transistor(T1) conduct in such a way as to cut off MOSFET (MF1). This leaves onlya small current flowing every half cycle of the AC main through theprogrammable Zener integrated circuit TL431 (Z2). The voltage togenerate the 15 volts of the 7815 regulator is generated from thevoltage over the motor by a suitable DC-DC converter.

Alternatively, the inductance L_(M) ((64) in FIG. 6) connected to theBLDCM can be replaced by a transformer, with one of its secondarywindings generating the voltage for the 7815.

I3. DC-DC Converter

FIG. 10C shows an implementation of the DC-DC converter necessary toextract useable voltage out of the motor voltage.

The DC-DC converter is designed to be controlled from the mainmicroprocessor. The design can be used to reduce or boost the voltagecollected from MOTOR_TOP, according to the mode of operation. Thecircuit can operate in three modes:

DCDC_m1: As a down converter switched mode power supply. The voltagefrom MOTOR_TOP is filtered by resistor (10C01) and capacitors (10C02)and (10C03). The transistor (10C04), diode (10C06) coil (10C05) andcapacitors (10C08) and (10C09) can operate as a down converter switchedmode power supply, when motor voltage is high. In this mode of operationthe MOSFET (10C12) is off and transistor (10C04) is controlled by apulse width modulator output from the microcontroller (PWM_PS). Themicroprocessor uses the voltage sampled by (10C11) (node labelledMPADPS) to close the loop. DCDC_m2: As a boost converter switched modepower supply. The voltage from MOTOR_TOP is filtered by resistor (10C01)and capacitors (10C02) and (10C03). Transistor (10C04) is saturated allthe time. The MOSFET (10C12), diode (10C07) coil (10C05) and capacitors(10C08) and (10C09) operate as a boost converter (also called a “ringingchoke”) switched mode power supply, when motor voltage is low. In thismode of operation, the MOSFET (10C12) is controlled by a pulse widthmodulator output from the microcontroller (PWM_PS). The microprocessoruses the voltage sampled by (10C11) (node labelled MPADPS) to close theloop. DCDC_m3: As a linear power supply. The voltage from MOTOR_TOP isfiltered by resistor (10C01) and capacitors (10C02) and (10C03).Transistor (10C04) is saturated all the time. The MOSFET (10C12) is offall the time. The coil (10C05) and capacitors (10C08) and (10C09) act asfurther filtering. Diode (10C07) conducts all the time.

The microprocessor selects the mode of operation based on the voltageover the capacitor (10C08), by operating the output line PS_MODE. Theline PS_MODE changes the way analog switches (10C14) and (10C13)connect.

I4. Start-up Switch

FIG. 10E shows an implementation of the switch (10D05) of FIG. 10D andthe circuit that operates the switch (10D06) in FIG. 10D. Whentransistor (10E13) is off, and the voltage at the input is lower thanthe Zener voltage of diode (10E10) plus the minimum voltage drop forconduction in the base of transistor (10E13), the transistor (10E03) isoff. The current taken from the input voltage is the leakage current ofall the transistors connected. When the voltage at the input is highenough for the base of (10E13) to conduct with current through the Zenerdiode (10E10), transistor (10E03) is turned on, as collector of (10E13)draws current and is pulled down. Also, transistor (10E06) is turned on.

The collector current of (10E06) goes through the base of (10E13)establishing a positive feedback loop, which keeps (10E13) conductingeven when the input voltage Vi becomes lower than the minimum voltagerequired to start the operation of the switch.

Transistor (10E04) and resistor (10E02) limit the current through thetransistor (10E03). In this way, large capacitors can be connected to Viand Vsw, without damaging the transistor (10E03).

I5. Flow Generator with Auxiliary Power Supply

FIG. 11A shows an airflow generator using the concept of auxiliary powersupply.

The topology is similar to that of FIG. 6 with the switch implemented byMOSFET (11A05), the DC power source being the rectified AC main as inFIG. 9A (note node VCCLINE). The inductance is coil (11A08) and thefreewheeling diode is (11A07). The capacitor in parallel with the BLDCMis (11A11). The current is sensed by resistor (11A14). The BLDCM is(11A12).

All the control circuit runs from +5 volts from a low voltage dropregulator (11A17). The auxiliary power supply (11A01) can be implementedwith the circuit of FIG. 10B, modified to give around 6 volts at theoutput rather than 12 volts. This is done by changing Zener diode (Z1)in FIG. 10B to a 5.6 volt Zener diode rather than an 11 volt one.

An alternative to the auxiliary power supply (11A01) is the arrangementof FIG. 10D. With the circuit of FIG. 10E replacing the switch (10D05)and the circuit block (10D06), the low voltage drop 5V regulator (11A17)and all the circuits running from 5 volts take the place of (10D12), andthe voltage adapter (10D13) is resistor (11A02) and capacitor (11A03).

The microcontroller (11A16) has inputs and outputs that can interfacewith a brushless DC motor and electronic commutator similar to the oneshown in FIG. 9. Consequently, the control lines out of (11A16) havebeen labelled consistently with the output lines out of (9J2). Thevoltage in the node MOTOR_TOP is monitored using an analog input to themicrocontroller (A/D converter) through network (11A15). The MOSFET(11A05) is driven by a pulse transformer (11A09). The electroniccommutator of the BLDCM can be pulse width modulated to enable a moreflexible operation. The effect is achieved by applying one of the PWMoutputs of the microcontroller to the gate (11A18).

FIG. 11B shows a current mode controller designed to work with thecircuit of FIG. 11A. One of the pulse width modulator output lines ofthe microcontroller clocks the “D” type flip-flop (11B05), setting theoutput to one. When the current reaches the level set by another pulsewidth modulator, through a filter formed by (11B12) and (11B13), thecomparator (11B06) resets the flip-flop. The output of the flip-flop isconnected to buffers that drive the transformer (11A09). Gate (11B04)and inverter (11B01) provide fail-safe operation. If the line labelledCDFF from the microcontroller is low, the output buffers are disabledand the flip-flop cannot be set to one.

The current is measured also by the microcontroller by reading the A/Dinput connected to the operational amplifier (11B11), which isamplifying the voltage sensed by the resistor (11A14). A crowbar circuitis shown in FIG. 11A (11A06) that is connected between MOTOR_TOP andMOTOR_BT as a protection device. The crowbar is a well-known circuit inwhich an SCR is triggered when the voltage across the terminals of thecircuit exceeds a given value, working as a protection againstmalfunctions. When the crowbar is triggered, the fuse (11A04) is blownto prevent further damage.

I6. Multiple Cooling Fan Controller

FIG. 12 shows another application of the embodiments of the invention,as disclosed in FIG. 7H. A transformer (1203) connected as in a flybackDC-DC converter is used to provide the inductance and the freewheelingdiode. In the circuit of FIG. 12, up to four 12 volt cooling fans can beconnected in series. This limit of four is to limit the maximum voltageat the output of the circuit. Each of the cooling fans is connectedthrough a connector and can be replaced by a link when not connected.

A current mode controller (1210) is used to control the MOSFET (1208)while sensing the current with resistor (1205).

The transformer has an additional winding to provide power to thecurrent mode controller through voltage regulator (1202). The circuitblock (1201) provides the power for starting the current modecontroller.

In the secondary side of the transformer, a microcontroller (1212) ispowered from voltage regulator (1205). (1205) takes the power from thevoltage resulting from the cooling fans when the set output current (Io)is circulating.

Resistor (1218) is used to sense the output current. The microcontroller(1212) can sense the voltages V0 to V3 in the circuit, corresponding tothe voltage across each of the cooling fans, and the top of the stack.Each fan has a capacitor in parallel, as shown in the block (1204).Temperature can be sensed by a sensor (1217) and read by an analog inputof the microcontroller. The microcontroller interacts with the currentmode controller through an isolated interface, as in FIGS. 9H and 9I, tochange the current (Io) and change the frequency of operation of thecurrent mode controller randomly to facilitate EMI compliance.

An isolated mains AC zero crossing circuit may be added to themicrocontroller, to allow for the current (Io) to be changed as a fullwave rectified AC signal synchronous with the AC main, in order toimprove the power factor of the circuit.

(1206) and (1207) are snubbers, which may be necessary in any flybackcircuit, depending on the characteristics of the transformer used.

I7. Cooling Fan

FIG. 13 shows a cooling fan based on a brushless DC motor running fromthe AC main with the system shown in FIG. 7A. A rectifier bridge (131)rectifies the AC main. The inductance (L_(M)), (135) corresponds withthe inductance (7A05) of FIG. 7A, while the capacitor in parallel withthe motor CM (134) corresponds with (7A03) of FIG. 7A. The anode of thefreewheeling diode (1350) is connected from the drain of power MOSFET(137) and the cathode of the freewheeling diode is connected to VCCLINE.

The network consisting of (1314), (1313) and (1315) provides power tostart a UCC3845 based current mode controller. Resistor (1322) sensesthe current through the switch (137). In operation, charge is “pumped”through the circuit consisting of capacitor (136), diodes (139) and(138) and Zener diode (1310). The voltage over the Zener diode (1310) isconnected through diode (1311) to capacitor (1312).

A microcontroller (1318) powered from a low voltage drop regulator(1323) is used to perform the following tasks:

Task_Fig13_1 Measure the voltage across the fan to check its operationand to have an indication of its speed. Task_Fig13_2 Set the currentthrough the cooling fan by Interfacing to the (1317). It can bemodulated to correct power factor by measuring the voltage VCCLINE andmaking the current through the coil to be a full wave AC rectifiedwaveform synchronous with the voltage in VCCLINE. If this is done,capacitor (132) is a small capacitor, allowing the voltage at the outputof the rectifier to track the AC line full wave rectified voltage.Task_Fig13_3 Change the frequency of operation of (1317) followingspread spectrum techniques to facilitate EMI regulatory compliance.(reference 8). Task_Fig13_4 Control the speed of the fan. Task_Fig13_5Interface through optocoupler (1321) to a temperature to frequencyconverter, to set the speed of the fan in response to changes in ambienttemperature.

The circuit of (1317) can be taken from FIGS. 9H and 9I, with the powertaken from capacitor (1312), through diode (1316) instead of the 12 voltregulator (9H2) of FIG. 9H.

I8. Bilevel Flow Generator with Power Factor Correction

FIG. 14 shows an application of an embodiment of the invention disclosedin FIG. 7I.

The AC main voltage is passed through a circuit similar to the circuitshown in FIG. 9A. The filtered AC mains voltage is rectified by bridge(141). Capacitor (142) is small enough for the voltage across it totrack the full wave rectified AC main.

Switch (1414) is a transformer driven power MOSFET, similar to theswitch of FIG. 11. Inductance (1415) has a function similar to (7I5),and a freewheeling diode (1420) is connected as (7I4) in FIG. 7I.

The motor driver (143) uses the pulse width modulation of the electroniccommutator to drive motor (144) in constant current mode, using sensor(146).

Controller (1412) controls the current through the inductance (1415),that is, approximately the current through the sensor (1417).

The current through the motor and the current through the inductance areset by the main controller (149).

Sensors (148) translate physical variables into information that (149)can read.

The system in FIG. 14 can operate in several modes as follows:

F14M_1. If the microcontroller measures the voltage in the pointconnecting C_(M) and L_(M), and use the measurement as feedback in acontrol loop that changes the current through the coil (1415) to obtainthe desired voltage over the motor, then the system operates as thecurrent state of the art. F14M_2. If the system operates as describedabove, but the voltage is changed to assist the action of the motordriver (143), the system is behaving in an innovative way. For instance,the voltage may be raised immediately before the motor needsacceleration. Or the voltage may change to make the action of the pulsewidth modulation more effective. F14M_3. The system can work asdisclosed in FIG. 7I. The average current that the motor controller(143) uses is allowed to pass through inductance (1415). Charge isaccumulated in C_(M), to be used when peak current is needed. F14M_4.This mode is similar to the previous mode (3), but the current throughinductance (1415), (L_(M)) is shaped as a full wave rectified sinusoidalwaveform, synchronous with the waveform in the rectified AC mains, toimprove power factor correction. This can be achieved either bymultiplying the current output by the value read in the analog todigital converter input through network (1410) or with the methoddescribed in with the description of FIG. 7I. The network (1411)connects the microcontroller (149) to the filtered AC mains, for zerocrossing detection. The result is shown immediately below (1411) in FIG.14.

Current sensor (146) does not need to be used, and the pulse widthmodulation of the microcontroller (149) can operate directly over theelectronic commutator of the BLDCM (144), in response to some physicalquantity that is controlled. Still, similar results can be achieved.

The feedback loop controlling the average current through the inductance(1414) has to use the voltage over the motor, to calculate the actualcurrent drawn by the motor itself. This is done by the calculation:Current_in_or_out_of_CM=CM*Diff_in_voltage/Time_between_samplingandI_motor=I_inductance−Current_in_or_out_of_CMwhere:

Diff_in_voltage: The change in capacitor voltage since the last sample.Current_in_or_out_of_CM: The net current through the terminal of thecapacitor. Time_between_sampling: The time between the samples made tothe voltage over the capacitor. CM: The value of the capacitor inparallel with the motor. I_motor: Average current through the motorduring sample interval. I_inductance: Current through the inductance ascontrolled. It is measured in the sensor (1417).

Current state of the art feedback loops, to control the voltage ofoutput of a power supply, use an error magnitude that would be thedifference between the voltage over the capacitor and the desiredvoltage. If the difference is large enough, the output of the controlloop will saturate.

In this new control scheme, even large voltage differences between thecapacitor instantaneous voltage and its average voltage do notnecessarily cause a similar response; the response depends on thecurrent requirements.

Table 2 lists pseudo-code following the syntax of the C languagedetailing the algorithm.

If the system also corrects for power factor, the sampling interval isone cycle of the AC main. The sampling interval can be made half thecycle if the extra harmonics introduced by occasionally having differentcurrent in both half cycles of the AC mains are acceptable.

TABLE 2 // algorithm to determine the value to set the current throughthe inductance (LM) // according to the charge stored in the capacitorin parallel with the motor (CM) VC =reading_capacitor_voltage_from_AD_converter() ; if ( VC > VCmax ) { I =0 ; } // check that the voltage is not too high else {   If ( VC = VCmin) { I = IMAX ; } // check that the voltage is not too low    else   {  V_Diff = VC− Old_VC ; // calculate the voltage difference betweensamples   // the two lines bellow are necessary if the average currentthrough the   // motor is unknown   I_cap = C * V_Diff /Time_between_samples ; // calculate the net current // into thecapacitor   I_motor = I_inductance − Icap ; // calculate the currentthough the motor   If ( VC > V_target )     {     // here if there is noneed for accumulating additional charge     I = I_motor ;     }   else    {  // here if the capacitor voltage is less than or equal to thetarget voltage     // calculate the current needed to replenish thecapacitor     I_to_charge_C = ( V_target − VC ) * C /Time_between_samples ;     // add the current needed to replenish to thecurrent needed by the motor     I = I_motor + I_to_charge_C ;     }   }} Old_VC= VC ; // save the current value of the capacitor voltage Return( I ) ; // return the calculated current // end of the function

Once the average current through the inductance is calculated, itremains constant for the sampling time. If the system is power factorcontrolled, the average current through the inductance is used tocalculate the peak of the full wave rectified sinusoidal current that isapplied through the inductance, synchronically with the AC mains as:I(t)=I_average through_inductance*π/2*|sin((2*π*F)*t)|where:

-   -   I(t) is the set current to the current mode controller.    -   I_average through_inductance is the current requirement keeping        the motor running and the capacitor with the target charge.    -   F is the frequency of the AC main.    -   t is the time.

The operating frequency of the current mode controller is changed usingthe algorithm shown in Table 3.

TABLE 3 Algorithm for changing the frequency randomly to improve EMIcompliance // Table of frequency data FT[0]= Data frequency 0; FT[1]=Data frequency 1; FT[2]= Data frequency 2; . . . . . . . . . . . . . .FT[N−1]= Data frequency N−1 ; Int Get_new_frequency(Old_frequency) {new_number: R = generate_random_number_between_N−1_and_0 ( ); If (Old_frequency = 0 ) and ( R=N−1 ) { goto new_number ; } If (Old_frequency = N−1 ) and ( R=0 ) { goto new_number ; } If ( R =Old_frequency ) { goto new_number ; } If ( R = Old_frequency − 1 ) or (R= Old_frequency +1 ) { goto new_number ; } Return ( R ) ; } // note thehardware is programmed with data from the table using the number asindex.FIG. 16—A Topology for an Active Power Factor Correcting Circuit

FIG. 16 shows a topology for an active power factor correcting circuit.Although related to other switch-mode power supply topologies, thecircuit is unique in the fact that the output is not referred to theinput ground, but rather to the input voltage.

The circuit could be described as “a non-isolated flyback switcher withthe output referred to the input”, or “a low side switched, buck-boostconverter with the output referred to the input”. It is also related tothe boost converter. The buck-boost converter is the switch-mode powersupply topology from which the circuit of FIG. 7L is derived. However,unlike the boost converter, the output voltage can be set higher orlower than the input voltage (the rectified AC mains).

In FIG. 16, a diode bridge (1601) rectifies the AC mains into a fullwave rectified AC waveform. The value of capacitor C_(SMALL) (1602) hasto be chosen so the voltage in the point “A” of the circuit (thepositive output of the diode bridge rectifier (1601)) tracks the voltageproduced by the rectifier. The output of the circuit of FIG. 16 is theport from “A” to V_(O), consequently the point “A” has been made theinput common terminal for the main converter (1607) in FIG. 16. The mainconverter (1607) is what is normally call the “load” of the circuit ofFIG. 16.

Capacitor Co (1606) is a storage capacitor. It is in parallel with theoutput. The switch (1605) in FIG. 16 is typically an N-Channel powerMOSFET. When the switch is closed, the inductive element L_(PFC) (1603)is connected across the input voltage and current flows into it, growinglinearly during the period of time the switch is conducting (T_(ON)). Atthe frequency of commutation normally used in this type of circuits, thefull wave rectified AC mains remains practically constant for the periodof commutation. The controller (1610) in FIG. 16 uses the informationprovided by the current sensor (1608) to decide when to open the switch(1605). When the switch (1605) opens, the diode (1604) takes the currentcirculating from the inductive element (1603). The current through theinductive element cannot change instantaneously and starts flowing intoboth the load and the storage capacitor Co (1606) connected in parallelwith the load. With the switch (1605) open and the diode (1604)conducting, the inductive element (1603) is just in parallel with theseries combination of the diode (1604) itself, in series with theparallel combination of Co (1606) and the load (main converter 1607). Ifthe switch is kept open, current keeps flowing, charging capacitor Co(the portion of the current not taken by the load) until the currentthough the inductive element reaches zero. At this moment, the diode(1604) becomes reverse-biased by the voltage across the storagecapacitor Co.

If the switch closes before the current reaches zero, then the circuitis said to operate in the continuous current mode. The process startsagain in the next cycle. By the way the Power Factor Controller (PFC)(1610) controls the maximum value of the current through the inductiveelement (1603), and the conduction time of the switch (1605), thefiltered waveform of the train of triangular (or trapezoidal for thecase of continuous current mode) current pulses taken from capacitor(1602) is filtered by the action of the capacitor into a normally goodapproximation of full wave rectified AC mains current waveformsynchronous with the AC mains, which yields close to unity power factorwhen “seen” from the AC mains side. The feedback circuit (1609) isneeded to provide the controller (1610) with the information of thevoltage across the storage capacitor Co (1606).

This topology can be seen as related to the system of FIG. 7J, which hasbeen analyzed in FIG. 15.

The analysis of the system in steady state is similar to the analysismade in FIG. 15. The “load” of the system is the block (1607). Any typeof DC-to-DC or DC-to-AC converter could be used in block (1607).Although FIG. 16 shows a converter, it will be obvious for those skilledin the art that any load suitable to be connected to a power supplycould be used in place of block (1607).

Consequently with the results of FIG. 15, it can be said that from adynamic point of view, the circuit can be designed using the well-knownequations of the buck-boost converter (also known as positive tonegative converter). However, it must be kept in mind that the outputrefers to the input voltage and not to ground, that the polarity of theoutput is not negative, and that the switch must be rated as the sum ofthe input voltage plus the output voltage.

The advantages of this topology are:

-   -   Low side switch (unlike the buck-boost or positive to negative        converter and the buck converter topology).    -   No in-rush current problem (unlike the boost converter).    -   High power factor correction (unlike the buck converter        topology).    -   Freedom to choose the output voltage. Normally it will be lower        than the output voltage to be able to simplify the converters        following the circuit. However the output voltage could be made        variable.

In the classical non-power-factor-corrected topology for an off themains switch-mode power supply implemented as a buck converter followedby a DC-DC converter operating from the lower voltage created by thebuck converter, it would be an advantage to replace the buck converterby the circuit of FIG. 16 to give power factor correction. However, asit would be clear for a designer skilled in the art, it must be notedthat for optimum performance the DC-DC converter following the circuitof FIG. 16 must be itself controlled with its own feedback loop from theoutput of the power supply.

The advantages of lowering the voltage obtained from the power factorcontroller are:

-   -   P type MOSFETs are rarely manufactured for drain to source        breakdown voltages over 200 volts. P channel MOSFETs permit more        flexible designs, with simple drivers.    -   MOSFETs with lower drain to source breakdown voltages may be        faster.    -   MOSFETs with lower drain to source breakdown voltages have lower        ON resistance.    -   Logic level MOSFETs are rarely manufactured for drain to source        breakdown voltages over 60 volts.    -   Working with lower voltages makes design and prototyping easier.    -   The voltages of many electronic circuits are very low. Hence, it        is an advantage to drive a high performance DC-DC converter from        a lower voltage.

For switched mode power supplies (DC-to-DC converter following thecircuit of FIG. 16) or other devices like fluorescent tubes drivers(DC-to-AC converter following the circuit of FIG. 16) the main converterfollowing the power factor corrector stage is isolated. Hence the factthat the output of the circuit of FIG. 16 is not referred to ground is aproblem only from the point of view of noise and for the complexity ofthe feedback circuit. Both problems can be overcome.

The disadvantage of not having the output referred to ground can beeasily overcome by decoupling carefully with capacitors with good highfrequency characteristics to form a ground plane at the frequency of thecommutation of the converter.

The feedback from the output of the controller has to be done by eitheran isolated link or using a differential amplifier. The most inexpensivetechnique would typically be to use an optocoupler. However, all thetechniques currently used for providing feedback though an isolatingbarrier can be used. Alternatively, a differential amplifier can be usedtaking the input from the two output terminals of the circuit of FIG.16. This eliminates the input voltage from the mains. This voltageappears as common mode voltage between the output terminals of thecircuit.

A topology based on a buck-boost converter may use a simple voltagedivider for the feedback circuit, and the output voltage can be lowerthan the input voltage. However, the requirement of a high side drivefor the switch far out-weighs the advantage of the simpler feedbackcircuit.

Any of the current state of the art techniques or algorithms for powerfactor correction can be used in the controller (1610) (see Ref. 10,page 222 or Ref. 25, chapter 1):

-   -   Fix-on time, Discontinuous Current Control (DCC), with or        without fixed output voltage (for the boost topology is called a        “boost follower”).    -   Critical Conduction Mode (CRM) also known as Transitional Mode        Controllers (Ref. 25, page 8).    -   Continuous Conduction Mode (CCM) Control (see Refs. 20, 21 and        25).

In continuous current control, the control loop can use as feedbackseveral alternatives, for instance average current control or peakcurrent control.

From a practical implementation point of view, most if not all thecurrent integrated circuits controllers designed forboost-converter-based power-factor-correction circuits can be used withlittle modification for the block (1610) of FIG. 16.

If the power factor correction controller is based on a microcomputerwith analog to digital converter the technique disclosed in thedescription of the controller of FIGS. 7I and 9 can be used.Alternatively, if the factor correction controller is based on amicrocomputer without analog to digital converter the algorithmdisclosed in TABLE 1 can be used.

If a classical controller with fix oscillation frequency, set byexternal components, is used along with a microcomputer, the techniqueshown in FIGS. 9H and 9I and FIG. 17F in combination with the algorithmof TABLE 3 can be used to change randomly the frequency of oscillationof the controller, improving on the EMI characteristics of the system.

FIG. 17—Application of Topology for Power Factor Corrected AC-to-DCConverter of FIG. 16

FIG. 17 shows a practical example of one possible implementation of thetopology shown in FIG. 16. The NCP1650 power factor controllerintegrated circuit, from On Semiconductors (see Ref. 26), has beenchosen for the example. The NCP1651 (see Ref. 27), from the samemanufacturer, is a more suitable device since it does not require theexternal start-up circuitry and the feedback input is specificallydesigned to work with the optically coupled circuit used in this type ofapplication. However, the NCP1650 has been selected instead of theNCP1651 to show that most if not all power factor correction integratedcircuits can be adapted to work with the topology of FIG. 16.

As with all the integrated circuits, its application must comply withthe design parameters defined by the manufacturer. The example of FIG.17 is a variation of the following application note from Onsemiconductor: AND8106/D, 100 Watt, Universal Input, PFC Converter. Itcan be found on page 67 of Reference 25. The differences with theoriginal application note are:

-   -   The topology of the circuit conforms to the topology of FIG. 16.        The original design in the application note is for a boost        converter. In the design of FIG. 17, the output voltage can be        lower than the peak voltage of the rectified AC mains or higher.        In contrast, the boost converter of the application note can        produce only a voltage higher than the rectified AC mains peak        voltage.    -   The design of FIG. 17 does not need in-rush current protection.        In contrast, a boost converter must charge the storage capacitor        to the peak of the rectified AC mains on power up and typically        needs in-rush current protection.    -   The output of the power factor controller in FIG. 16 does not        share the same common terminal with the rectified AC mains or        the integrated circuit NCP1650. In FIG. 17F there is a reference        list of the four different common terminals used in the circuit        of FIG. 17.    -   The start-up circuit (FIG. 17B) has been modified because the        output voltage is not referred to the ground of the integrated        circuit.    -   The feedback circuit (FIG. 17C) has been modified because the        output voltage is not referred to the ground of the integrated        circuit.    -   The circuit in FIG. 17F is optional. If the circuit is added, a        microcontroller can produce pseudo-random variations in the        frequency of the oscillator of the NCP1650 integrated circuit.        This is another example of the idea disclosed in FIG. 91.    -   A generic DC-DC converter is added in FIG. 17E. This is an        example of the block (1607) in FIG. 16.

In FIG. 17A, a diode bridge (1701) rectifies the AC mains into a fullwave rectified AC waveform. The value of capacitor C1 (1702) has beenchosen so the voltage in the point V₁ of the circuit (the positiveoutput of the diode bridge rectifier (1701)) tracks the voltage producedby the rectifier. The circuit also includes capacitor (1711) that hasbeen placed in another side of the picture to show the fact thatcapacitor (1602) in FIG. 16 can be made of a number of capacitors inparallel. The position of the capacitors has to be considered withattention to EMI compatibility of the final product. All the capacitorsin parallel with C1 (1702) (or (1602) in FIG. 16) must have good highfrequency characteristics. It must be noted that the output of thecircuit of FIG. 17A is the port from V_(I) to V_(OP). Consequently thepoint V_(I) has been made the common for the DC-DC converter circuit inFIG. 17E.

Capacitor C3 (1707) is the storage capacitor of the power factorcontroller. It is in parallel with the output. The MOSFET MF1 (1709) inFIG. 17A is the switch (1605) of FIG. 16. When the MOSFET is conducting,the inductance L1 (1703) is connected across the input voltage andcurrent flows into it, growing linearly during the period of time theMOSFET is conducting (T_(ON)). At the frequency of commutation of thecircuit, the full wave rectified AC mains remains practically constantfor the period of commutation. The IC controller in FIG. 17, the NCP1650of FIG. 17D, is not referenced to the negative terminal of the diodebridge. Instead it is referred to the mid-point between the MOSFET(1709) and the current sensing resistor R2 (1708) in FIG. 17A. The chipuses the negative voltage with respect to the common node of the chip'scircuit (pin 15 in FIG. 17D) as current information to decide when toturn off the MOSFET MF1 (1709).

When the MOSFET (1709) is off, diode D2 (1706) takes the currentcirculating from the inductance (1703). The current through theinductance cannot change instantaneously and starts flowing into boththe load and the storage capacitor C3 (1707) connected in parallel withthe load, with the MOSFET MF1 non-conducting and the diode D2conducting. The inductance is just in parallel with the seriescombination of the diode D2, itself in series with the parallelcombination of the capacitor C3 (1707) and the load (converter of FIG.17E). If the MOSFET is kept non-conducting, current keeps flowing,charging capacitor C3 (the portion not taken by the load) until itreaches zero and the diode D2 becomes reverse-biased from the voltageacross the storage capacitor C3.

If the MOSFET conducts before the current reaches zero, then the circuitis said to operate in the continuous current mode. The process startsagain in the next cycle. By the way the integrated circuit NCP1650(1733) controls the maximum value of the current through the inductance,and the conduction time of the MOSFET, the filtered waveform of thetrain of triangular current pulses (or trapezoidal, in continuouscurrent mode) taken from capacitor C1 (1702) is filtered by the actionof the capacitor into a normally good approximation of full waverectified AC mains current waveform synchronous with the AC mains, whichyields close to unity power factor when “seen” from the AC mains side.

The inductance L1 has an additional winding that provides power for thechip itself. The inductance L1 works as the primary of a transformer.The diode D1 (1704) and the capacitor C2 (1705) provide rectificationand filtering of the wave in the secondary side of the arrangement.

FIG. 17B is the start-up sub-circuit. The circuit is similar to theapplication note referenced above. The changes reflect the fact that theoutput cannot bias the MOSFET to turn it off. Hence capacitors (1719)and (1720) have been added. Also added are resistor (1715) dischargecapacitors (1719) and (1720) when the system is not powered. MOSFET MF2(1717) is made to conduct because of the voltage difference betweenV_(BIAS) (that is low, at the start-up) and the voltage over the zenerdiode Z1 (1718). As V_(BIAS) approaches the working voltage thedifference with the zener value is smaller and the MOSFET (1717) stopsconducting. As the input of the start-up circuit is the full waverectified AC mains (and not the output storage capacitor as in theoriginal application note), without the capacitors (1719) and (1720) thevoltage over the zener diode is pulsating (when the input voltage of thefull wave rectified AC mains is lower than the zener value) and theMOSFET is periodically turned on. Hence without the capacitors (1719)and (1720), the start-up circuit overheats.

FIG. 17C is the feedback path. Those skilled in the art of switched modepower supplies will recognize the standard optically coupled feedbacknetwork formed with IC2 (1730), typically a TL431 programmable zenerdiode IC. Resistors network (1731) formed by R15, RV1 and R14 sets thevalue of the output voltage. When the output voltage grows, the voltagedrop across resistor R13 (1732) also grows. If this happens, the currentthrough the diode of the optocoupler IC3 (1729) grows too. The outputtransistor of the optocoupler conducts more and the voltage overresistor R11 (1728) grows. This voltage is directly fed to the input ofthe error comparator of the NCP1650 control loop (pin 6 of (1733)). Thenetwork formed by resistors R8 (1725), R9 (1726) and diode (1727) hasbeen added because the feedback network is not taken from a groundreferenced storage capacitor as in the original application not boostconverter. The network formed by resistor R8 (1725), R9 (1726) and diode(1727) guarantees that there is a minimum voltage in the input of theerror comparator of the NCP1650 control loop (pin 6 of (1733)) when thereference voltage of the device is enabled after the power-up sequencefinishes. The network (1724) is for compensating the feedback loop. Thevalues depend on the type of load and the specifications of theconverter.

FIG. 17D is the controller IC itself, the NCP1650 (1733). All thecomponents in the circuit are mandatory as per the manufacturerspecifications. Refer to the data sheet of the device (Ref. 26) and itsapplication notes (page 67 of Ref. 25).

FIG. 17D shows an example, in block diagram of the block (1607) in FIG.16. A classical push-pull converter has been chosen as an example only.Any type of DC-DC or DC-to-AC converter could be used in block (1607).Detailed operation and design methodologies of controller (17E02),MOSFETs (17E03) and (17E04), all components coupled with transformer(17E05), the output filtering network (7E06 to 17E10) and thetransformer (17E05) itself, can be found in the following references:

Reference 3, page 220.

Reference 7, pages 116 and 153.

Reference 9, page 37.

Reference 10, pages 2.147 to 2.151 and 2.153 to 2.159.

Reference 11, pages 34-38.

The feedback network for the converter in FIG. 17E is based on the samecomponents and operating principle as the feedback network used in FIG.17C. However, components R04, C03 and C02 (network 17E16) have beenadded for completeness. The network (17E16) is used for dynamiccompensation of the feedback loop. Inmost of the circuits of this type,compensation is added around IC5 (17E13) (unlike however the NCP1650that is designed so the compensation is added to a pin 7 of the chip(1724) in FIG. 17C). The graphic in FIG. 17E shows the full waverectified AC mains in the common point of the circuit in the primaryside of transformer (17E05). The reference point of the secondary sidecan be chosen independently of any other point in the circuit (providethat the isolation given by the specifications of the transformer isadequate). In FIG. 17E, earth has been chosen for the common point (or“ground terminal”) of the output of converter (1607).

FIG. 17F shows a list of the four different common terminals symbolsused in the circuit. FIG. 17F also shows an example of how theoscillator of NCP1650 can be frequency modulated by an external circuit.A microcontroller producing a pulse width modulated waveform in thediode terminals of the optocoupler IC06 (17F08) can producepseudo-random variations in the frequency of the oscillator of theNCP1650 integrated circuit. This is another example of the ideadisclosed in FIG. 91.

When current flows though the LED of the optocoupler IC06 (17F08) theoutput transistor conducts and the pulse width modulated waveform isreproduced over the resistor R105 (17F10). Resistor R104 (17F09) is usedto drop the voltage of the VREF supply by forming a resistive dividerwith resistor R105 (17F10). Resistor R103 (17F07) and capacitor C100(17F06) filter the pulse width modulated signal and its low frequencycontent is available in the non-inverting input of the operationalamplifier IC5 (17F04). The operational amplifier IC5 (17F04), itsfeedback network (R101 (17F03), R100 (17F01)) and the P channel MOSFETMF5 (17F02) form a regulated current source in parallel with the currentsource of 200 micro-amperes provided by pin 14 of the CP1650 IC, tocharge the timing capacitor CT (1738). Resistor R102 (17F05) is includedfor stability. If R101 is made equal to several hundred times the valueof R100, the current out of the drain terminal of the P channel MOSFETMF5 (17F02) will be the voltage difference between V_(REF) and thevoltage at the non-inverting input of the op-amp IC5 (17F04) divided bythe value of resistance R100 (17F01). In this way, the additionalcurrent charging the timing capacitor CT (1738) will be varied with thelow frequency content of the pulse width modulated signal at the inputof the opto-coupler.

While only a limited number of embodiments have been disclosed, numerousmodifications and substitutions can be made without departing from thescope and spirit of the invention.

REFERENCES

-   1. Brush-less Permanent Magnet Motor Design

Duane C. Hanselman

McGraw Hill Inc 1994

ISBN 0-07-026025-7

-   2. A250 Watt Current—Controlled SMPS with Synchronous Rectification

By R. Pearce and D. Grant

Application note 960A. International Rectifier

Page 137 of

HEXFET Designer's Manual Volume I,

HDM-1 first printing, International rectifier 1993

-   3. Power Electronics

Converters, Applications, and Design

N. Mohan, T. M. Undeland and W. P. Robbins

John Wiley & Sons 1989

ISBN 0-471-61342-8

-   4. Motorola Application Note AN-876

Using Power MOSFETs in Stepping Motor Control

Published in the proceedings of Powercon 9, 1982

-   5. UC1637/2637/3637 Switched Mode Controller for DC Motor Drive data    sheet

Unitrode Integrated Circuits Corp. (Texas Instruments Incorporated)

-   6. UC1637/2637/3637 Switched Mode Controller for DC Motor Drive

Application note U-102

Unitrode Integrated Circuits Corp. (Texas Instruments Incorporated)

7. Design of Solid States Power Supplies

Third Edition

Hnatek, E. R.

Van Nostrand Reinhold Co. 1989

ISBN 0-442-20768-9

-   8. EMC for Product Designers

Meeting the European EMC Directive

Tim Williams

Butterworth Heinemann 1992

ISBN 0-7506-1264-9

9. Switching Power Supply Design

Abraham I. Pressman

McGraw-Hill, Inc 1991

ISBN 0-07-050806-2

-   10. Switch Mode Power Supply Handbook

Keith Billings

McGraw-Hill, Inc 1989

ISBN 0-07-005330-8

-   11. Practical Switching Power Supply Design

Marty Brown

Academic Press Inc 1990

ISBN 0-12-137030-5

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Marty Brown

Butterworth Heinemann 2001

ISBN 0-7506-7329-X

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SGS-Thomson Microelectronics Application Note

AN282/0589

Page 305, Designer's Guide to Power Products Application Manual

2^(nd) Edition, June 1992

SGS-Thomson Microelectronics

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Page 29/42, FIG. 36 (use as a motor speed controller)

SGS-Thomson Microelectronics Application Note

AN244/1288

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SGS-Thomson Microelectronics 2^(nd) Edition, June 1992

-   15. How to Drive DC Motors With Smart Power IC's

By Herbert Sax

SGS-Thomson Microelectronics Application Note

AN 380/0591

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-   16. Load Current Sensing in Switch-mode Bridge Motor Driving    Circuits

By Herbert Sax

SGS-Thomson Microelectronics Application Note

AN 452/0392

Page 231 of Designer's Guide to Power Products Application Manual

SGS-Thomson Microelectronics 2^(nd) Edition, June 1992

-   17. Driving DC Motors

By Maiocchi

SGS-Thomson Microelectronics Application Note

AN 281/0189

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-   18. Switched-mode Drives for DC Motors

By Lester J. Hadley, Jr.

Philips Semiconductors Corporation, Application note

AN1221, December 1988

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L292 Integrated circuit data sheet

SGS-Thomson Microelectronics, March 1993

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Page 4/194 of 1994 Linear Data Book Volume III

Linear Technology Corporation

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By B. Wilkinson and J. Mandelcorn

U.S. Pat. No. 4,677,366

Jun. 30, 1987

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Publication Order Number: MC33035/D, April 2001-Rev. 4

Page 3050 in:

Analog Integrated Circuits, DL128/D

Power Management, Signal Conditioning and ASSP Devices

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Document Number HBD853/D Rev. 1, June 2004

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On Semiconductor

Document Number NCP1650/D Rev. 8, August 2003

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On Semiconductor

Document Number NCP1651/D Rev. 5, October 2003

1. A system for driving a direct-current (DC) motor under conditions ofcontrolled DC current, from a DC voltage source of a value larger thansad a motor operating voltage, independently of the operating voltage ofsaid motor, said system comprising: a pair of nodes for connection ofsaid DC motor, said nodes to be referred to herein as the first node andthe second node, said second node being connected to a common electricalterminal of the system through an electrical path with low impedance,including low impedance to DC current, said DC motor being connectedbetween said first node and said second node; an inductive element tostore energy and to act as a current source for said DC motor, saidinductive element being external to said DC motor, and not part of amain magnetic circuit of said DC motor, said inductive element beingconnected to said first node, in series with said DC motor, saidinductive element being capable of operating in a buck converter at thepower level required to operate said DC motor and at the frequency ofcommutation of a first switch, the terminal of said inductive elementnot connected to said first node being connected to a third node; saidfirst switch being connected to said inductive element at the thirdnode, the terminal of said inductive element remote from said DC motor,said first switch for connecting and disconnecting said inductiveelement to a direct current (DC) voltage source, a terminal of said DCvoltage source not connected to said first switch, being connected tosaid common electrical terminal of the system; said first switch being acontrolled switch capable of being turned off and on by control signalsfrom a control system, said control system operating based on an errorsignal and a value of the desired operating current for said DC motorset externally to the system, said control system turning said firstswitch off and on to minimize said error signal and to keep theoperating current of said DC motor at said desired value, said firstswitch being a single pole switch; a second switch connected betweensaid third node and said common electrical terminal of the system,parallel with a combination of said inductive element and said DC motorarranged in series, said second switch being controlled so that acurrent circulating through said inductive element circulates throughsaid second switch if said first switch is turned off and disconnectssaid inductive element from said DC voltage source, said second switchbeing a single pole switch; a capacitor arranged for connection inparallel with said DC motor to limit a resulting voltage over said DCmotor, said capacitor being capable of operating in a buck converter ata power level required to operate said DC motor and at the frequency ofcommutation of said first switch, said capacitor being connected betweensaid first node and a low impedance path to said common electricalterminal of the circuit; a current sensor for measuring a currentthrough said DC motor, the output of said current sensor being connectedto said control system of said first switch to generate said errorsignal for the operation of said control system controlling theoperation of said first switch; and means for controlling operation ofsaid second switch dependent upon the state of the first switch.
 2. Thesystem according to claim 1, wherein the voltage of said DC voltagesource is larger than the nominal rated voltage of said DC motor.
 3. Thesystem according to claim 1, wherein said second switch is a diodeconnected with appropriate polarity so that current circulating throughsaid inductive element circulates through said diode and if said firstswitch is open, disconnecting said inductive element.
 4. The systemaccording to claim 1, wherein said second switch is an electronic switchfor synchronous rectification connected with appropriate polarity sothat current circulating through said inductive element circulatesthrough said electronic switch and if said first switch is open,disconnecting said inductive element.
 5. The system according to claim1, wherein said first switch is an electronic switch.
 6. The systemaccording to claim 1, wherein said inductive element is an inductor, ora winding of a transformer.
 7. The system according to claim 1, whereina current through the inductive element is controlled independently froma current through the motor, the balance of electrical charge beingaccumulated or taken from the capacitor in parallel with the motor. 8.The system according to claim 7, wherein the current through the motoris calculated from the variation of the voltage across the capacitor inparallel with the motor.
 9. The system according to claim 7, whereinsaid DC motor is an electronically driven motor, including a brushlessmotor, said motor being braked electronically so that a current producedduring the braking process further charges said capacitor in parallelwith said motor.
 10. The system according to claim 1, wherein a currentthrough the inductive element is modulated as a full wave rectifiedsinusoid synchronous with the AC main voltage so that the power factorof the system, as a load to the AC main is improved.
 11. The systemaccording to claim 10, wherein the instant in which the sinusoidalwaveform of the AC main crosses zero is sensed to synchronize themodulation performed to the current through the inductive element withthe waveform in the AC main.
 12. The system according to claim 1,wherein said DC motor includes a brushless DC motor.
 13. The systemaccording to claim 12, wherein said DC motor includes an electroniccommutator for said brush less brushless DC motor.
 14. The systemaccording to claim 1, further comprising means for calculating saidcurrent through the motor dependent upon current measured in anotherpart of said system.
 15. The system according to claim 1, wherein afrequency of a pulse width modulated waveform, resulting from operationof said switches, is randomized to facilitate EMI compliance.
 16. Thesystem according to claim 1, wherein the voltage over the DC motor isused to estimate the speed of the motor.
 17. The system according toclaim 1, wherein said first switch and said direct current (DC) voltagesource are implemented by an electronic system that connects saidinductive element either to a given voltage or to a high impedance for aperiod of time determined by said control system controlling said firstswitch, said high impedance being measured with respect to said commonelectrical terminal of said system, said given voltage being setexternally to said system, said electronic system being connected tosaid common electrical terminal of the system, said electronic systembeing controlled by said control system controlling said first switch.18. A system for driving a direct-current (DC) motor under conditions ofcontrolled DC current, independently of the operating voltage of saidmotor, said system comprising: a pair of nodes for connection of said DCmotor, said nodes to be referred to herein as the first node and thesecond node, said second node being connected to a common electricalterminal of the system through an electrical path with low impedance,including low impedance of DC current, said DC motor is being connectedbetween said first node and said second node; a capacitor arranged forconnection in parallel with said motor to limit a resulting voltage oversaid motor, one terminal of said capacitor being connected to said firstnode, the other terminal of said capacitor to be connected through a lowimpedance to said common terminal of the system, said capacitor beingcapable of operating in a buck converter at the power level required tooperate said DC motor and at the frequency of commutation of a firstswitch; an inductive element with one terminal being connected to saidcommon terminal of the system through a low impedance path, the otherterminal of said inductive element, referred to herein as the thirdnode, being connected to said first switch, said inductive element beingused to store energy and to act as a current source for said DC motor,said inductive element being external to said DC motor and not part of amain magnetic circuit of said DC motor, said inductive element beingcapable of operating in a buck converter at the power level required tooperate said DC motor and at the frequency of commutation of said firstswitch; said first switch being connected to said inductive element inthe third node, said first switch for connecting and disconnecting saidthird node to a DC voltage source, a terminal of said DC voltage sourcenot connected to said first switch, being connected to said commonelectrical terminal of the system, said first switch being a controlledswitch capable of being turned off and on by control signals from acontrol system, said control system operating based on an error signaland a value of a desired operating current for said DC motor, setexternally to the system, said control system for turning said firstswitch off and on to minimize said error signal and to keep theoperating current of said DC motor at said desired value, said firstswitch being a single pole switch; a second switch being connectedbetween said first node and said third node, in series with the parallelcombination of said motor and said capacitor, and being connected to thecommon node between the first switch and said inductive element, saidsecond switch being controlled so that a current circulating throughsaid inductive element circulates through said second switch if thefirst switch is turned off and disconnects the third node from said DCvoltage source, said second switch being a single pole switch; a currentsensor for measuring a current through said DC motor, the output of saidcurrent sensor being connected to said control system of said firstswitch to generate said error signal for the operation of said controlsystem, controlling the operation of said first switch; and means forcontrolling operation of said second switch dependent upon the state ofthe first switch.
 19. The system according to claim 18, wherein saidfirst switch is an electronic switch, and said second switch is a diodeconnected with appropriate polarity so that current circulating throughsaid inductive element circulates through said diode and if said firstswitch is open, disconnecting said inductive element.
 20. The systemaccording to claim 18, wherein said inductive element is an inductor, ora winding of a transformer.
 21. The system according to claim 18,wherein a current through the inductive element is controlledindependently from a current through the motor, the balance ofelectrical charge being accumulated or taken from the capacitor inparallel with the motor.
 22. The system according to claim 21, whereinsaid DC motor is an electronically driven motor, including a brushlessmotor, said motor being braked electronically so that a current producedduring the braking process further charges said capacitor in parallelwith said motor.
 23. The system according to claim 18, wherein a currentthrough the inductive element is modulated as a full wave rectifiedsinusoid synchronous with the AC mains voltage so that the power factorof the system, as a load to the AC main is improved.
 24. The systemaccording to claim 18, wherein said first switch and said direct current(DC) voltage source are implemented by an electronic system thatconnects said inductive element either to a given voltage or to a highimpedance for a period of time determined by said control systemcontrolling said first switch, said high impedance being measured withrespect to said common electrical terminal of said system, said givenvoltage being set externally to said system, said electronic systembeing connected to said common electrical terminal of the system, saidelectronic system being controlled by said control system controllingsaid first switch.
 25. The system according to claim 18, wherein saidinductive element is an inductor and said second switch is connected toa terminal of said inductor that is not connected to said third node orto said common electrical terminal of the system.
 26. A system fordriving a direct-current (DC) motor under conditions of controlled DCcurrent, independently of the operating voltage of said motor, saidsystem comprising: a pair of nodes for connection of said DC motor, saidnodes to be referred to herein as the first node and the second node,said DC motor being connected between said first and said second node; acapacitor arranged for connection in parallel with said motor, betweensaid first node and said second node, to limit a resulting voltage oversaid motor, said first node, connected to a terminal of said capacitorand said motor, being also connected to a DC voltage source, saidcapacitor being capable of operating in a buck converter at the powerlevel required to operate said DC motor and at the frequency ofcommutation of a first switch, the other terminal of said DC voltagesource being connected to a common electrical terminal of the system; aninductive element with one terminal connected to said first node, acommon node of said DC voltage source, said capacitor and said DC motor,the other terminal of said inductive element, referred to herein as thethird node and being connected to said first switch, said inductiveelement being used to store energy and to act as a current source forsaid DC motor, said inductive element being external to said DC motor,and not part of a main magnetic circuit of said DC motor, said inductiveelement being capable of operating in a buck converter at the powerlevel required to operate said DC motor and at the frequency ofcommutation of said first switch; said first switch being connected tosaid inductive element in the third node, the other terminal of saidfirst switch, not connected to the third node being connected to saidcommon electrical terminal of the system through an electrical path withlow impedance, including low impedance to DC current, said first switchfor connecting and disconnecting the third node to said commonelectrical terminal of the system, said first switch being a controlledswitch capable of being turned off and on by control signals from acontrol system, said control system operating based on an error signaland a value of a desired operating current for said DC motor setexternally to the system, said control system for turning said firstswitch off and on to minimize said error signal and to keep theoperating current of said DC motor at said desired value, said firstswitch being a single pole switch; a second switch being connectedbetween said second and said third node, said second switch beingcontrolled so that a current circulating through said inductive elementcirculates through said second switch if the first switch is turned offand disconnects the third node from said common electrical terminal ofthe system, said second switch being a single pole switch; a currentsensor for measuring a current through said DC motor, the output of saidcurrent sensor being connected to said control system of said firstswitch to generate said error signal for the operation of said controlsystem, controlling the operation of said first switch; and means forcontrolling operation of said second switch dependent upon the state ofthe first switch.
 27. The system according to claim 26, wherein saidfirst switch is an electronic switch, and said second switch is a diodeconnected with appropriate polarity so that current circulating throughsaid inductive element circulates through said diode and if said firstswitch is open, disconnecting said inductive element.
 28. The systemaccording to claim 26, wherein said inductive element is an inductor, ora winding of a transformer.
 29. The system according to claim 26,wherein a current through the inductive element is controlledindependently from a current through the motor, the balance ofelectrical charge being accumulated or taken from the capacitor inparallel with the motor.
 30. The system according to claim 29, whereinsaid DC motor is an electronically driven motor, including a brushlessmotor, said motor being braked electronically so that a current producedduring the braking process further charges said capacitor in parallelwith said motor.
 31. The system according to claim 26, wherein a currentthrough the inductive element is modulated as a full wave rectifiedsinusoid synchronous with the AC mains voltage so that the power factorof the system, as a load to the AC main is improved.
 32. The systemaccording to claim 26, wherein said first switch and said direct current(DC) voltage source are implemented by an electronic system thatconnects said inductive element either to a given voltage or to a highimpedance for a period of time being measured with respect to saidcommon electrical terminal of said system, said given voltage being setexternally to said system, said electronic system being connected tosaid common electrical terminal of the system, said electronic systembeing controlled by said control system controlling said first switch.33. The system according to claim 26, wherein said inductive element isan inductor and said second switch is connected to a terminal of saidinductor that is not connected to said third node or to said directcurrent (DC) voltage source.
 34. A system for driving a direct-current(DC) motor under conditions of controlled DC current, from a DC voltagesource of a value larger than a motor operating voltage, independentlyof the operating voltage of said motor, said system comprising: a pairof nodes for connection of said DC motor, said nodes referred to as thefirst node and the second node, said second node being connected to adirect current (DC) voltage source, a terminal of said DC voltage sourcenot connected to said second node being connected to a common electricalterminal of the system through an electrical path with low impedance,including low impedance to DC current, said DC motor being connectedbetween said first node and said second node; an inductive element tostore energy and to act as a current source for said DC motor, saidinductive element being external to said DC motor and not part of a mainmagnetic circuit of said DC motor, said inductive element beingconnected to said first node in series with said DC motor, saidinductive element being capable of operating in a buck converter at thepower level required to operate said DC motor and at the frequency ofcommutation of a first switch, the terminal of said inductive elementnot connected to said first node being connected to a third node; saidfirst switch being connected to said inductive element at said thirdnode, the terminal of said inductive element remote from said DC motor,said first switch for connecting and disconnecting said inductiveelement to said common electrical terminal of the system, a terminal ofsaid first switch not connected to said third node being connected tosaid common electrical terminal of the system, said first switch being acontrolled switch capable of being turned off and on by control signalsfrom a control system, said control system operating based on an errorsignal and a value of the desired operating current for said DC motorset externally to the system, said control system turning said firstswitch off and on to minimize said error signal and to keep theoperating current of said DC motor at said desired value, said firstswitch being a single pole switch; a second switch connected betweensaid third node and said second node, parallel with a combination ofsaid inductive element and said DC motor arranged in series, said secondswitch being controlled so that a current circulating through saidinductive element circulates through said second switch if said firstswitch is turned off and disconnects said inductive element from saidcommon electrical terminal of the system, said second switch being asingle pole switch; a capacitor arranged for connection in parallel withsaid DC motor to limit a resulting voltage over said DC motor, saidcapacitor being capable of operating in a buck converter at a powerlevel required to operate said DC motor and at the frequency ofcommutation of said first switch, said capacitor being connected betweensaid first node and said second node; a current sensor for measuring acurrent through said DC motor, the output of said current sensor beingconnected to said control system of said first switch to generate saiderror signal for the operation of said control system controlling theoperation of said first switch; and means for controlling operation ofsaid second switch dependent upon the state of the first switch.
 35. Thesystem according to claim 34, wherein said first switch is an electronicswitch, and said second switch is a diode connected with appropriatepolarity so that current circulating through said inductive elementcirculates through said diode and if said first switch is open,disconnecting said inductive element.
 36. The system according to claim34, wherein said inductive element is an inductor, or a winding of atransformer.
 37. The system according to claim 34, wherein a currentthrough the inductive element is controlled independently from a currentthrough the motor, the balance of electrical charge being accumulated ortaken from the capacitor in parallel with the motor.
 38. The systemaccording to claim 37, wherein said DC motor is an electronically drivenmotor, including a brushless motor, said motor being brakedelectronically so that a current produced during the braking processfurther charges said capacitor in parallel with said motor.
 39. Thesystem according to claim 34, wherein a current through the inductiveelement is modulated as a full wave rectified sinusoid synchronous withthe AC mains voltage so that the power factor of the system, as a loadto the AC main is improved.
 40. The system according to claim 34,wherein said first switch and said direct current (DC) voltage sourceare implemented by an electronic system that connects said inductiveelement either to a given voltage or to a high impedance for a period oftime being measured with respect to said common electrical terminal ofsaid system, said given voltage is being set externally to said system,said electronic system being connected to said common electricalterminal of the system, said electronic system is being controlled bysaid control system controlling said first switch.
 41. A system fordriving a direct-current (DC) motor under conditions of controlled DCcurrent, from a DC voltage source of a value smaller than a motoroperating voltage, independently of the operating voltage of said motor,said system comprising: a pair of nodes for connection of said DC motor,said nodes to be referred to herein as the first node and the secondnode, said DC motor being connected between said first and said secondnode; a capacitor arranged for connection in parallel with said motor,between said first node and said second node, to limit a resultingvoltage over said motor, said first node being connected to a commonelectrical terminal of the system through an electrical path with lowimpedance, including low impedance to DC current, said capacitor beingcapable of operating in a buck converter at the power level required tooperate said DC motor and at the frequency of commutation of a firstswitch; wherein said direct current (DC) voltage source has one terminalof said being connected to an inductive element, the other terminal ofsaid DC voltage source being connected to said common electricalterminal of the system; said inductive element with one terminalconnected to said DC voltage source, the other terminal of saidinductive element, referred to herein as the third node, being connectedto said first switch, said inductive element being used to store energyand to act as a current source for said DC motor, said inductive elementbeing external to said DC motor, and not part of a main magnetic circuitof said DC motor, said inductive element being capable of operating in abuck converter at the power level required to operate said DC motor andat the frequency of commutation of said first switch; said first switchbeing connected to said inductive element in the third node, the otherterminal of said first switch, not connected to the third node beingconnected to said common electrical terminal of the system through anelectrical path with low impedance, including low impedance to DCcurrent, said first switch for connecting and disconnecting the thirdnode to said common electrical terminal of the system, said first switchbeing a controlled switch capable of being turned off and on by controlsignals from a control system, said control system operating based on anerror signal and a value of a desired operating current for said DCmotor set externally to the system, said control system for turning saidfirst switch off and on to minimize said error signal and to keep theoperating current of said DC motor at said desired value, said firstswitch being a single pole switch; a second switch connected betweensaid second and said third node, said second switch being controlled sothat a current circulating through said inductive element circulatesthrough said second switch if the first switch is turned off anddisconnects the third node from said common electrical terminal of thesystem, said second switch being a single pole switch; a current sensorfor measuring a current through said DC motor, the output of saidcurrent sensor being connected to said control system of said firstswitch to generate said error signal for the operation of said controlsystem, controlling the operation of said first switch; and means forcontrolling operation of said second switch dependent upon the state ofthe first switch.
 42. The system according to claim 41, wherein saidfirst switch is an electronic switch, and said second switch is a diodeconnected with appropriate polarity so that current circulating throughsaid inductive element circulates through said diode and if said firstswitch is open, disconnecting said inductive element.
 43. The systemaccording to claim 41, wherein said inductive element is an inductor, ora winding of a transformer.
 44. The system according to claim 41,wherein a current through the inductive element is controlledindependently from a current through the motor, the balance ofelectrical charge being accumulated or taken from the capacitor inparallel with the motor.
 45. The system according to claim 44, whereinsaid DC motor is an electronically driven motor, including a brushlessmotor, said motor being braked electronically so that a current producedduring the braking process further charges said capacitor in parallelwith said motor.
 46. The system according to claim 41, wherein a currentthrough the inductive element is modulated as a full wave rectifiedsinusoid synchronous with the AC mains voltage so that the power factorof the system, as a load to the AC main is improved.
 47. The systemaccording to claim 41, wherein said first switch and said direct current(DC) voltage source are implemented by an electronic system thatconnects said inductive element either to a given voltage or to a highimpedance for a period of time determined by said control systemcontrolling said first switch, said high impedance being measured withrespect to said common electrical terminal of said system, said givenvoltage being set externally to said system, said electronic systembeing connected to said common electrical terminal of the system, saidelectronic system being controlled by said control system controllingsaid first switch.
 48. The system according to claim 41, wherein saidinductive element is an inductor and said second switch is connected toa terminal of said inductor that is not connected to said third node orto said direct current (DC) voltage source.